Semiconductor device

ABSTRACT

A semiconductor device includes: a voltage-control-type clock generation circuit having a plurality of stages of first delay elements and whose oscillation frequency is controlled according to a control voltage applied to the first delay elements; a delay circuit having a plurality of stages of second delay elements connected serially; and a selection circuit selecting one from pulse signals output by the plurality of stages of respective second delay elements. The first delay elements and the second delay elements have a same structure formed on a same semiconductor substrate, and a delay amount of the second delay elements is adjusted according to the control voltage.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a division of and claims the benefit of priorityunder 35 U.S.C. §120 from U.S. Ser. No. 13/110,971 filed May 19, 2011,which is a divisional of U.S. Ser. No. 12/986,500 filed Jan. 7, 2011(now U.S. Pat. No. 7,973,580 issued Jul. 5, 2011), which is a divisionalof U.S. Ser. No. 12/351,426 filed Jan. 9, 2009 (now U.S. Pat. No.7,893,744 issued Feb. 22, 2011), and claims the benefit of priorityunder 35 U.S.C. §119 from Japanese Patent Application Nos. 2008-117491filed Apr. 28, 2008, Japanese Patent Application No. 2008-004894 filedJan. 11, 2008 and Japanese Patent Application No. 2008-298025 filed Nov.21, 2008; the entire contents of each of which are incorporated hereinby reference.

BACKGROUND OF THE INVENTION

This invention relates to a semiconductor device in which a digitalcontrol power source is formed.

For example, PWM (Pulse Width Modulation) control in an analog powersource is performed by comparison of triangle wave and feedback voltage(for example, Patent document 1 (JP-A 2001-251370 (Kokai)). In thiscase, a pulse width realized by analog PWM circuit can continuouslychange.

On the other hand, in a PWM circuit in a digital power source, time canbe merely set discretely. When the PWM control is performed by using aclock, the clock of 1 GHz becomes required for realizing a timeresolution of 1 nanosecond, and the clock of 10 GHz becomes required forrealizing a time resolution of 100 picoseconds. In forming a circuitgenerating a clock having such a high frequency on a semiconductorsubstrate, a most-advanced process is required, and there is a problemthat the consumption current increases for operation by the clock.

Moreover, conventionally, a plurality of power sources are connected inparallel. For example, when ten power sources having an output currentof 10 ampere are operated in parallel, a power source having an outputcurrent of 100 ampere can be composed.

A current digital power IC having a parallel running function is “MasterControl Architecture” composed of one master IC controlling switching ofall of the phases and a plurality of driver ICs. A problem thereof isthat the power system becomes failed in itself if a trouble is caused inthe master IC from any cause.

Moreover, in Non-patent document 1 (“Current Sharing in DigitallyControlled Masterless Multi-phase DC-DC Converters”, Power ElectronicsSpecialists Conference, 2005. PESC '05. IEEE 36th), a masterlessarchitecture in which a plurality of ICs each having capability ofbecoming the master are connected in parallel to realize parallelrunning has been disclosed. However, a master clock is input from theoutside, and the current information is shared among the power ICs (thephases) through a high-speed digital bath, and therefore, the number ofterminals is large and the power consumption is also large.

Moreover, reference voltage used for controlling the output voltage toconverge to be a target value is generalized (shared) among the powersource ICs. This leads to increase of the number of the terminals inactual productization, and the reference voltage-shared terminals areaffected by, wiring resistance on the substrate, parasitic capacitance,and noise, and therefore, it is necessary to care for wiring among thepower ICs and the design is troublesome.

Moreover, in the case of multi-phase operation, its interleaving settingis performed by setting outside the power ICs, and if a trouble iscaused in any one of the phases from any cause, the power system becomesfailed in itself, and therefore, the advantage of the masterlessarchitecture has been lost.

SUMMARY OF THE INVENTION

According to a first aspect of the invention, there is provided asemiconductor device including: a voltage-control-type clock generationcircuit having a plurality of stages of first delay elements and whoseoscillation frequency is controlled according to a control voltageapplied to the first delay elements; a delay circuit having a pluralityof stages of second delay elements connected serially; and a selectioncircuit selecting one from pulse signals output by the plurality ofstages of respective second delay elements, the first delay elements andthe second delay elements having a same structure formed on a samesemiconductor substrate, and a delay amount of the second delay elementsbeing adjusted according to the control voltage.

According to a second aspect of the invention, there is provided asemiconductor device including: a voltage-control-type clock generationcircuit having a plurality of stages of first delay elements and whoseoscillation frequency is controlled according to a first control voltageapplied to the first delay elements; a delay circuit having a pluralityof stages of second delay elements connected serially; a selectioncircuit selecting one from pulse signals output by the plurality ofstages of respective second delay elements; an error detection circuitdetecting disagreement of a delay amount in the delay circuit; and anerror adjustment circuit applying a second control voltage that is thefirst control voltage compensated based on the detection result of theerror detection circuit, the first delay elements and the second delayelements having a same structure formed on a same semiconductorsubstrate, and a delay amount of the second delay elements beingadjusted according to the second control voltage.

According to a third aspect of the invention, there is provided asemiconductor device including a digital PWM circuit outputting: a firstpulse signal switching from a low level to a high level at a timing ofbeing synchronized with a clock signal and switching from the high levelto the low level at a timing set by a shorter time resolution than thatof a period of the clock signal; and a second pulse signal switchingfrom a low level to a high level at a timing set by a shorter timeresolution than that of a period of the clock signal and switching fromthe high level to the low level at a timing set by a shorter timeresolution than that of a period of the clock signal, the digital PWMcircuit including: a delay circuit having a plurality of stages of delayelements connected serially; a first selection circuit selecting onefrom outputs of the respective delay elements and outputting a signalsetting a falling timing of the first pulse signal; a second selectioncircuit selecting one from outputs of the respective delay elements andoutputting a signal setting a rising timing of the second pulse signal;and a third selection circuit selecting one from outputs of therespective delay elements and outputting a signal setting a fallingtiming of the second pulse signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic cross-sectional view showing the circuit structureformed in a semiconductor device according to a first embodiment of theinvention;

FIG. 2 is a circuit diagram of a first delay element shown in FIG. 1;

FIGS. 3A to 3C are waveform charts of main signals in avoltage-control-type clock generation circuit shown in FIG. 1;

FIGS. 4A to 4F are timing charts of the main signals in the circuitshown in FIG. 1;

FIG. 5 is a schematic view showing the circuit structure formed in asemiconductor device according to a second embodiment of this invention;

FIG. 6 is a schematic view showing the circuit structure formed on asemiconductor device according to a third embodiment of the invention;

FIG. 7 is a schematic view showing the structure of an error detectioncircuit in FIG. 6;

FIGS. 8A to 8C are timing charts for each signal va, ve1, ve2 shown inFIG. 6, FIG. 8A shows the case where the delay amount varies to thedirection of becoming slower, FIG. 8B shows the case where the delayamount varies to the direction of becoming smaller, and FIG. 8C showsthe case where the delay amount does not vary;

FIG. 9 is a schematic view showing the structure of an error adjustmentcircuit in FIG. 6;

FIG. 10 is a schematic view showing the circuit structure formed on asemiconductor device according to a fourth embodiment of the invention;

FIG. 11 is a schematic view showing the structure of the semiconductordevice according to this embodiment of the invention based on acompletely differential circuit as the delay element;

FIG. 12 is a circuit diagram of the delay element shown in FIG. 11;

FIG. 13 is a schematic view showing a modification example without usingEXOR circuit in the clock generation circuit shown in FIG. 11;

FIG. 14 is a schematic view showing the structure of a voltage step downDC-DC converter as one example of a digital control power sourceaccording to this embodiment of the invention;

FIG. 15 is a circuit diagram showing one structure example of a digitalPWM circuit shown in FIG. 14 as a semiconductor device according to afifth embodiment of the invention;

FIGS. 16A to 16K are waveform charts (timing charts) of the main signalsin the circuit shown in FIG. 15;

FIG. 17 is a circuit diagram showing another structure example of thedigital PWM circuit shown in FIG. 14 as a semiconductor device accordingto a sixth embodiment of the invention;

FIG. 18 is a circuit diagram showing still another structure example ofthe digital PWM circuit shown in FIG. 14 as a semiconductor deviceaccording to a seventh embodiment of the invention;

FIG. 19 is a circuit diagram showing still another structure example ofthe digital PWM circuit shown in FIG. 14 as a semiconductor deviceaccording to an eighth embodiment of the invention;

FIG. 20 is a circuit diagram showing still another structure example ofthe digital PWM circuit shown in FIG. 14 as a semiconductor deviceaccording to a ninth embodiment of the invention;

FIG. 21 is a circuit diagram showing one structure example of an outputvoltage control circuit in a DC-DC converter as a semiconductor deviceaccording to an twelfth embodiment of the invention;

FIG. 22 is a schematic view showing one example of thresholds (Bin) setin comparing the reference voltage Vref with the output voltage in thecircuit in FIG. 21;

FIG. 23 is a schematic view showing the corresponding relation betweenthe thresholds shown in FIG. 22 and the control parameter set inaccordance with this;

FIG. 24 is a circuit diagram showing one structure example of an outputvoltage control circuit in a DC-DC converter as a semiconductor deviceaccording to an thirteenth embodiment of the invention;

FIG. 25 is a schematic view showing one example of thresholds (Bin) setin comparing the reference voltage Vref with the output voltage in thecircuit in FIG. 24;

FIG. 26 is a schematic view showing the corresponding relation betweenthe thresholds shown in FIG. 25 and the control parameter set inaccordance with this;

FIG. 27 is a schematic view showing the structure of a digital controlpower source as a semiconductor device according to a fourteenthembodiment of the invention;

FIGS. 28A and 28B are schematic views for describing the multiphaseoperation in a parallel operation power source;

FIG. 29 is a graph showing the relation between a current flowingthrough each of power ICs and the efficiency in a digital control powersource as a semiconductor device according to a fifteenth embodiment ofthe invention;

FIG. 30 is a schematic view showing the structure of a digital controlpower source as a semiconductor device according to a sixteenthembodiment of the invention;

FIG. 31 is a schematic view showing the structure of a digital controlpower source as a semiconductor device according to a seventeenthembodiment of the invention;

FIG. 32 is a schematic view showing a specific example for atransmit/receive circuit of share current value information provided oneach power source IC shown in FIG. 31;

FIG. 33 is a waveform chart showing an example which each IC powersource IC shown in FIG. 31 converts its own output current value into ahigh-level pulse width;

FIG. 34 is a waveform chart for describing a wired OR function in thecurrent share structure shown in FIG. 31;

FIG. 35 is a waveform chart showing an example which each IC powersource IC shown in FIG. 31 converts its own output current value into alow-level pulse width;

FIG. 36 is a schematic view showing a structure example of each DC-DCconverter shown in FIG. 31;

FIGS. 37A to 37G are waveform charts (timing chart) of the main signalsin the circuit of FIG. 19;

FIG. 38 is a circuit diagram showing a structure example of a digitalPWM circuit in a semiconductor device according to a tenth embodiment ofthe invention;

FIGS. 39A to 39D are waveform charts (timing chart) of the main signalsin the circuit of FIG. 38; and

FIG. 40 is a circuit diagram showing a structure example of a digitalPWM circuit in a semiconductor device according to a eleventh embodimentof the invention.

DETAILED DESCRIPTION OF THE INVENTION

Hereinafter, embodiments of this invention will be described withreference to drawings.

First Embodiment

FIG. 1 is a schematic cross-sectional view showing a circuit structureformed in a semiconductor device according to a first embodiment of theinvention.

The circuit shown in FIG. 1 is a digital PWM (Pulse Width Modulation)circuit and has a voltage-control-type clock generation circuit 10, adelay circuit 20, a delay element input circuit 32, a counter 31, amultiplexer 33, a flipflop 34, and so forth. These are formed on acommon semiconductor substrate and composed as one semiconductor device(chip or packaged shape).

The voltage-control-type clock generation circuit 10 has delay elementsA0, A1, A(n+1)/2, . . . An (the suffix n is odd number) serving as aplurality of stages of first delay elements. This circuit has a ringoscillator structure in which (n+1) delay elements A0, A1, A(n+1)/2, . .. An are cascade-arranged and the output of the last-stage delay elementAn is input to an inverter circuit 70 and the output of the invertercircuit 70 is input to the first-stage delay element A0.

An example of the circuit structure of the delay elements A0, A1,A(n+1)/2, . . . An and the inverter circuit 70 is shown in FIG. 2.

Each of the delay elements A0, A1, A(n+1)/2, . . . An has a structure inwhich two inverter circuits each composed of a p-type MOS transistor Mp2and an n-type MOS transistor Mn1 are cascade-arranged. The gates of thetransistor Mp2 and the transistor Mn1 are connected to each other, andthe drain in which the transistors Mp2, Mn1 are connected to each otherserves as the output terminal, and this output terminal is connected tothe gate in which the next-stage transistors Mp2, Mn1 are connected toeach other.

Furthermore, a p-type MOS transistor Mp1 serving as a transistorcontrolling the operation current is connected between a power wire of apower voltage VDD and the transistor Mp2, and an n-type MOS transistorMn2 is connected between the transistor Mn1 and a ground node. Theoperation current of the transistor Mp1 is controlled by a bias voltageVb1. The operation current of the transistor Mn2 is controlled by a biasvoltage Vb2. The inverter 80 being capable of controlling the operationcurrent outputs a value inverted to the input. Therefore, in order tooutput the delayed signal without inverting the signal in each of thedelay elements A0, A1, A(n+1)/2, two inverter circuits 80 arecascade-arranged.

The inverter circuit 70 connected to the output terminal of thelast-stage delay element An has only one structure that is the same asthe inverter 80 composed of the p-type MOS transistor Mp2 and the n-typeMOS transistor Mn1. In order that the ring oscillator oscillates, it isnecessary that the stage number of the inverter circuits 80 is odd.Accordingly, by inputting the output of the last-stage delay element Anto the inverter circuit 70 and inputting the output of the invertercircuit 70 to the first-stage delay element A0, the stage number is setto be odd.

By referring to FIG. 1 again, the output of the first-stage delayelement A0 (shown in FIG. 3( a)) and the output of the delay elementA(n+1)/2 (shown in FIG. 3( b)) are input to an exclusive disjunction orEXOR (EXclusive OR) circuit 11, and the EXOR circuit 11 output a clocksignal clk shown in FIG. 3( c). Thereby, the clock generation circuit 10can output a signal clk having a frequency that is two-times larger thana frequency oscillated by the plurality of delay elements A0, A1,A(n+1)/2, . . . , An and the inverter circuit 70.

In the present embodiment, the clock signal clk is generated by usingthe EXOR circuit 11, but another circuit except for the EXOR circuit ispossible as long as outputting the frequency that is two-times largerthan the frequency oscillated by the plurality of delay elements A0, A1,A(n+1)/2, . . . An, and the inverter circuit 70.

The frequency clock signal clk depends on the delay amount (delay time)of each of the delay elements A0, A1, A(n+1)/2, . . . An, and the delayamount depends on the current flowing through each of the delay elementsA0, A1, A(n+1)/2, . . . An. And, the current flowing each of the delayelements A0, A1, A(n+1)/2, . . . An is controlled by the control voltageVsrc. That is, in the voltage-control-type clock generation circuit 10,the oscillation frequency is controlled according to the applied controlvoltage Vsrc.

The delay circuit 20 also has delay elements B0, B1, B2, . . . Bn (thesuffix n is odd) serving as a plurality of stages of second delayelements that are cascade-arranged in the same manner as the clockgeneration circuit 10, and the delay elements B0, B1, B2, . . . Bn areconnected not in a ring shape but linearly. That is, the output of thelast-stage delay element Bn is not returned to the input side of thefirst-stage delay element B0.

The delay elements B0, B1, B2, . . . Bn in the delay circuit 20 have thesame circuit structure as the delay elements A0, A1, A(n+1)/2, . . . Anin the clock generation circuit 10 so as to be synchronized with theclock frequency (frequency of the clock signal clk). That is, the delayelements A0, A1, A(n+1)/2, . . . An, and the delay elements B0, B1, B2,. . . Bn in the delay circuit 20 are the delay elements having the samestructures formed on the same semiconductor substrate (semiconductorchip), and each of the delay elements B0, B1, B2, . . . Bn also have thesame circuit structures as the delay elements A0, A1, A(n+1)/2, . . . Andescribed above with reference to FIG. 2.

And, the current flowing through each of the delay elements B0, B1, B2,. . . Bn is controlled by the same control voltage Vsrc as applied tothe clock generation circuit 10, and therefore, the delay amount of eachof the delay elements B0, B1, B2, . . . Bn is adjusted by the controlvoltage Vsrc.

In the former stage of the delay circuit 20, the delay element inputcircuit 32 is provided, and to the delay element input circuit 32, acount value cnt of the counter 31 and, for example, a digital signal D[MSB] with top 3 bits of a 5-bit digital signal are input.

Here, FIG. 4 is a timing chart of the main signals clk, cnt, S, q0, q1,q2, q3, R, and V0 in the circuits of FIG. 1.

The counter 31 counts up the clock signals clk oscillated by the clockgeneration circuit 10 one by one (by one period) as shown in FIG. 4( b),and when the counted value (3 bits) cnt accords to D[MSB], the delayelement input circuit 32 outputs a pulse signal va to the first-stagedelay element B0. The pulse signal va is a pulse signal having a widthof one period of the clock signal clk.

The example shown in FIG. 4 indicates the case of “011”, and the pulsesignal va is input to the first-stage delay element B0 at the timingthat the count value cnt becomes “011”, and the pulse signal vatransmits through each of the delay elements B0, B1, B2, . . . Bnsequentially from the first stage, and delay according to the controlvoltage Vsrc is generated as shown in FIG. 4( d) in the timing that eachof the pulse signals q0, q1, q2, q3, . . . output by the respectivedelay elements B0, B1, B2, . . . Bn rises.

Each of the output q0, q1, q2, . . . of the respective delay elementsB0, B1, B2, . . . Bn is input, for example, to the multiplexer 33 as aselection circuit. The multiplexer 33 selects one of the output q0, q1,q2, . . . of the respective delay elements B0, B1, B2, . . . Bn on thebasis of D[LSB] with low 2 bits in the 5-bit digital signal whose top 3bits are used as the above-described D[MSB], and the selected output isoutput to the reset terminal of the flipflop 34 as the reset pulse R asshown in FIG. 4( e). The example shown in FIG. 4 indicates the case thatthe D[LSB] has “10”, and the output q2 is selected, and the reset pulseR synchronized with the q2 is output to the reset terminal of theflipflop 34.

To the set terminal of the flipflop 34, the set pulse S shown in FIG. 4(c) is input. The set pulse S has a predetermined period, and becomes ON,for example, when the count value cnt of the counter 31 is “000”.

The flipflop 34 outputs a pulse signal V0 from the output terminal. Asshown in FIG. 4( f), the pulse signal V0 switches from a low level to ahigh level at the rising edge of the set pulse S, and the high level isheld until the reset pulse R is input, and when the reset pulse R isinput, the high level switches to the low level at the rising of thereset pulse R.

This pulse signal V0 becomes an output of the PWM circuit shown in FIG.1, and the pulse signal V0 is provided to the gate of the switchingelement (MOSFET), ON/OFF of the switching element is controlled.

According to this embodiment, based on the above-described set pulse Sand the reset pulse R, PWM control of the pulse signal V0 is performed.The set pulse S rises based on the count value cnt of counting up theclock signal clk one by one (period by period), and by rising of the setpulse S, the pulse signal V0 becomes in the high level from the lowlevel. Therefore, the rising edge of the pulse V0 is relatively roughlydetermined according to what number of the clock clk the edge is in.

By contrast, the falling edge of the pulse signal V0 is determined by arising edge of any one of the outputs q0, q1, q2, q3, . . . of therespective delay elements B0, B1, B2, . . . Bn. Each of the rising edgeof q0, q1, q2, q3, . . . is delayed with respect to the former stage bysmaller time interval than one period of the clock clk, and therefore, apulse width modulation of the pulse signal V0 is realized by smallertime resolution than one period of the clock clk according to which(rising edge of) signal is selected out of q0, q1, q2, q3, . . . .Thereby, without requiring high-speed clock, the pulse width modulationcan be realized by smaller time resolution, and increase of const orconsumption current can be suppressed.

The delay elements A0, A1, A2, . . . An of the clock generation circuit10 and the delay elements B0, B1, B2, B3, . . . Bn of the delay circuit20 are composed by the same circuits formed on the same semiconductorsubstrate, and the delay amounts thereof are controlled by the samecontrol voltage Vsrc, and therefore, resolution of accurately dividingthe clock period can be simply obtained. In the circuit structure ofFIG. 1, one clock period can be divided into about (n+1). For example,in the case of dividing it into 32, n=31 is set, and the clock signalclk is generated by the output of the delay element A0 and the output ofthe delay element A16. As the clock period becomes longer, thetransmission delay in the delay circuit 20 becomes larger, andconversely, if the clock period becomes shorter, the transmission delayin the delay circuit 20 becomes shorter.

Second Embodiment

FIG. 5 is a schematic view showing a circuit structure formed in asemiconductor device according to a second embodiment of the invention.The same signs are appended to the same components as theabove-described first embodiment, and the detailed explanation thereofwill be omitted.

In the second embodiment, the structure of the clock generation circuitis different from that of the first embodiment.

The clock generation circuit 15 in the second embodiment is formed bythe delay elements A0, A1, . . . A(n+1)/2 that are half of those of theclock generation circuit 10 and the inverter circuit 70, and the clocksignal clk is output from the output terminal of the inverter circuit70. Moreover, the EXOR circuit is used in generating the clock signalclk in the first embodiment, but is not required in the secondembodiment. Thereby, in this embodiment, the number of the delayelements of the clock generation circuit 15 decreases, and thereby, theoccupied area in the semiconductor chip can be reduced compared to thefirst embodiment.

Third Embodiment

FIG. 6 is a schematic view showing a circuit structure formed on asemiconductor device according to a third embodiment of the invention.The same signs are appended to the same components as theabove-described first embodiment, and the detailed explanation will beomitted.

Each of the clock generation circuit 10 and the delay circuit 25 havethe delay elements formed on the same semiconductor substrate by thesame process, and the same control voltage Vsrc is applied thereto, andtherefore, in principle, by the rising edges of the outputs q0, q1, q2,in the delay circuit 25, one period of the clock clk is almostaccurately evenly divided (delay amount per stage becomes 1/(number ofthe stages) of the clock period), but if the pair properties of thedelay elements in both of the circuits 10, 25 are broken, the risingedges of the outputs q0, q1, q2, come not to accurately evenly dividethe clock period, and disagreement of the delay amount such as progressor lag of the delay occasionally caused.

Accordingly, in this embodiment, in addition of the above-describedstructure of the first embodiment, an error detection circuit 42 and anerror adjustment circuit 41 are provided as shown in FIG. 6. Moreover,in this embodiment, a delay element B(n+1) is further additionallyconnected to a latter stage of the last-stage delay element Bn. Thedelay element B(n+1) is also formed on the same semiconductor substrateand has the same circuit structure, as the delay elements B0, B1, Bn.

The error detection circuit 42 detects disagreement of the delay amountin the delay circuit 25 on the basis of, a signal va input to thefirst-stage delay element B0, an output signal ve2 of the delay elementB(n+1) that is a latter stage of the last stage, and an output signalve1 of the last-stage delay element Bn.

In this embodiment, in order to perform the control so that the risingedge of the output signal ve1 of the last-stage delay element Bn iscontained accurately in one clock period, the delay element B(n+1) isfurther added at the latter stage of the last-stage delay element Bn.

If the rising edge of the output signal ve1 of the last-stage delayelement Bn is not accurately contained in one clock period, thefollowing thing is generated.

The case of linearly inputting a command increasing a value “1” by “1”is thought. The value D[MSB] is fixed and D[LSB] increases “1” by “1”,and the nodes are selected sequentially so as to monotonously increaselike q0, q1, . . . qn. When the node of qn is output, ideally, the delaytime becomes equal to one clock period. Then, in D[LSB], “+1” is addedto the value of D[MSB]. In this case, in the circuit, the external oneclock period+the delay time of the delay element B0 are output. In thiscase, if the output of the node of qn becomes longer than one clockperiod, subsequently, the time output by one clock period+delay elementB0 occasionally becomes short according to the delay amount of the delayelement B0 despite increasing the value, there is possibility that thelinearity is broken and that the control is largely affected.Accordingly, in this embodiment, in order to perform the control so thatthe rising edge of the output signal ve1 of the last-stage delay elementBn is contained accurately in one clock period, the delay element B(n+1)is further added.

The structure of the error detection circuit 42 is shown in FIG. 7.

The output signal ve1 of the last-stage delay element Bn is input to theinput terminal of the flipflop 42 a, and the output signal ve2 of thedelay element B(n+1) of the latter stage of the last stage is input tothe input terminal of the flipflop 42 b. And, the input signal va to thefirst-stage delay element B0 is input to each of the flipflops 42 a, 42b at the rising edge as a set pulse.

The signal va is a pulse having a width of one period of the clock clk,and as shown in FIG. 8, by the falling edge of the pulse va, each of theoutput signal ve1 of the last-stage delay element Bn and the outputsignal ve2 of the delay element B(n+1) that is the latter stage of thelast stage is latched.

When the delay amount varies to the direction of becoming slower, bothof ve1, ve2 are latched to the low level by the falling edge of thesignal va as shown in FIG. 8A, and the value of “00” is output to theerror adjustment circuit 41. In this case, the error adjustment circuit41 increases the current flowing through the delay elements B0, B1, . .. B(n+1). That is, when the current flowing through the delay elementsA0, A1, A(n+1)/2, . . . An of the clock generation circuit 10 is Isrc1and the current Isrc2 flowing through the delay elements B0, B1, . . .B(n+1) of the delay circuit 20 is Isrc2, Isrc2 is increased more thanIsrc1 (Isrc2=Isrc1+Δi). Because the operation current increases, thedelay elements B0, B1, . . . B(n+1) operates at a higher speed than thecase of adding Δi. Accordingly, feedback is generated to the directionof making the delay amount smaller.

By contrast, when the delay amount varies to the direction of becomingsmaller, both of ve1, ve2 are latched to the high level by the fallingedge of the signal va as shown in FIG. 8B, and the value of “11” isoutput to the error adjustment circuit 41. In this case, the erroradjustment circuit 41 decreases Isrc2 more than Isrc1 (Isrc2=Isrc1−Δi).Because the operation current decreases, the delay elements B0, B1, . .. B(n+1) operates at a lower speed than the case of reducing Δi.Accordingly, feedback is generated to the direction of making the delayamount larger.

When ve1 is latched to the high level and ve2 is latched to the lowlevel at the falling edge of the signal va as shown in FIG. 8C, thedelay amount is determined not to have disagreement, the value of “10”is output to the error adjustment circuit 41. In this case, the erroradjustment circuit 41 does not increase and decrease Isrc2, but thecurrent Isrc2 is held.

When the detection result by the error detection circuit 42 is “00” or“11”, the error adjustment circuit 41 adjusts Isrc2 so that “00” or “11”becomes “10”. That is, the error adjustment circuit 41 applies thesecond control voltage Vsrc2 that is the compensated first controlvoltage Vsrc1 applied for flowing the current Isrc1 through the delayelements A0, A1, A(n+1)/2, . . . An, to the delay circuit 25, andthereby, the current Isrc2 flows through the delay elements B0, B1, . .. B(n+1) of the delay circuit 25 by the second control voltage Vsrc2.

One example of the circuit structure of the error adjustment circuit 41is shown in FIG. 9.

The error adjustment circuit 41 has a decoder 43, for example eightswitching elements (N-type MOS) M1 to M8 ON/OFF-controlled withreceiving the signal from the decoder 43 to the gates thereof, and acurrent mirror circuit composed of P-type MOSs 51, 53 and N-type MOSs52, 54.

In the initial state, all of the output ports i1 to i8 of the decoder 43output “0”, and all of the eight switching elements M1 to M8 are set tobe in the OFF state, and accordingly, the current does not flow throughthe lines L1 and L2. In this case, P-type MOS 51 is set to be in the ONstate by the first control voltage Vsrc1 applied to the gate thereof,and thereby the same current flows through the nodes n1, n2. The biasvoltage corresponding to this current is applied to the gate of theP-type MOS 55 corresponding to each of the delay elements B0, B1, B2, .. . B(n+1), and the P-type MOS 55 becomes in the ON state, and thecurrent is supplied to each of the delay elements B0, B1, B2, . . .B(n+1).

When the detection result in the above-described error detection circuit42 is “00”, first, “1” is output only from the port i5 (the other portsi1 to i4 and i6 to i8 remain “0”). When the port i5 becomes “1”, theswitching element M5 becomes ON, the current flows through the line L2.When the current flowing through the line L2 is I1 and the currentflowing through the node n2 in the above-described initial state is I,the current of I+I1 flows through the node n2. And, the gate of theP-type MOS 55 is biased according to the current of and as a result, alarger current than that of the initial state flows through each of thedelay elements B0, B1, B2, . . . B(n+1), and the delay amount iscompensated to the direction of becoming smaller.

After the compensation, if the detection result in the error detectioncircuit 42 still remains “00”, “1” is also output from the port i6 inaddition of the port i5. When the ports i5 and i6 become “1”, theswitching elements M5 and M6 become ON, the current two-times largerthan I1 flows through the line L2. Accordingly, the current of I+(2×I1)flows through the node n2, and the gate of the P-type MOS 55 is biasedaccording to the current of I+(2×I1), and the current flowing througheach of the delay elements B0, B1, B2, . . . B(n+1) is adjusted to belarger, and the delay amount is compensated to the direction of becomingsmaller.

When the detection result in the error detection circuit 42 becomes“10”, the above-described current adjustment in the error adjustmentcircuit 41 is finished, but if the detection result in the errordetection circuit 42 still indicates “00”, by sequentially setting theports i7, i8 to be “1”, the current flowing through each of the delayelements B0, B1, B2, . . . B(n+1) is adjusted so as to be larger.

When the detection result in the above-described error detection circuit42 is “11”, first, “1” is output only from the port i1 (the other portsi2 to i4 and i5 to i8 remain “0”). When the port i1 becomes “1”, theswitching element M1 becomes ON, the current flows through the line L1.When the current flowing through the line L1 is I1 and the currentflowing through the node n1 in the above-described initial state is I,the current of I-I1 flows through the node n1. And, the gate of theP-type MOS 55 is biased according to the current of I-I1, and as aresult, a smaller current than that of the initial state flows througheach of the delay elements B0, B1, B2, . . . B(n+1), and the delayamount is compensated to the direction of becoming slower.

After the compensation, if the detection result in the error detectioncircuit 42 still remains “11”, “1” is also output from the port i2 inaddition of the port i1. When the ports i1 and i2 become “1”, theswitching elements M1 and M2 become ON, the current two-times largerthan 11 flows through the line L1. Accordingly, the current of I-(2×I1)flows through the node n1, and the gate of the P-type MOS 55 is biasedaccording to the current of I-(2×I1), and the current flowing througheach of the delay elements B0, B1, B2, . . . B(n+1) is adjusted to besmaller, and the delay amount is compensated to the direction ofbecoming slower.

When the detection result in the error detection circuit 42 becomes“10”, the above-described current adjustment in the error adjustmentcircuit 41 is finished, but if the detection result in the errordetection circuit 42 still indicates “11”, by sequentially setting theports i3, i4 to be “1”, the current flowing through each of the delayelements B0, B1, B2, . . . B(n+1) is adjusted so as to be smaller.

As described above, by adjusting the current flowing through each of thedelay elements B0, B1, B2, . . . B(n+1) on the basis of the detecteddisagreement of the delay amount to compensate the disagreement of thedelay amount, it can be realized that one period of the clock clk isalmost accurately evenly divided by rising edges of the outputs q0, q1,q2, . . . (delay amount per stage becomes 1/(number of the stages) ofthe clock period). As a result, the precise and reliability of the PWMcontrol can be improved.

In the above-described specific example, four-stage adjustment can beperformed for both of the directions of making the delay smaller andslower, but here, the assumed disagreement of the delay amount is onlybased on variation of the semiconductor process, and considering thecurrent semiconductor process, the disagreement of the delay amountcannot be thought to be too large, and the disagreement can besufficiently addressed by the adjustment widths of several stages(several bits) as shown in FIG. 9, and thereby, cost up beyond necessitycan be suppressed.

Fourth Embodiment

FIG. 10 is a schematic view showing a circuit structure formed on asemiconductor device according to a fourth embodiment of the invention.The same signs are appended to the same components as theabove-described embodiments, and the detailed explanation will beomitted.

In the fourth embodiment, the structure of the clock generation circuitis different from that of the third embodiment. The clock generationcircuit 15 in the fourth embodiment is formed by the delay elements A0,A1, . . . A(n+1)/2 that are half of those of the clock generationcircuit 10 in the third embodiment and the inverter circuit 70, and theclock signal clk is output from the output terminal of the invertercircuit 70. Moreover, the EXOR circuit 11 is used in generating theclock signal clk in the third embodiment, but is not required in thefourth embodiment. Thereby, in this embodiment, the number of the delayelements of the clock generation circuit 15 decreases, and thereby, theoccupied area in the semiconductor chip can be reduced compared to thethird embodiment.

In the above-described embodiments, one-input-and-one-output circuit isused as the delay element, but, a completely differential circuit may beused as shown in FIGS. 11, 12.

In FIG. 11, the clock generation circuit 16 has (n+1) delay elements C0,C1, . . . Cn. In the above-described embodiments, the inverter circuit70 is connected to the latter stage of the last-stage delay element, andby contrast, in the structure shown in FIG. 11, without using theinverter 70, the ring oscillator structure is formed by (n+1) delayelements C0, C1, . . . Cn.

In each of the first-stage C0 to (n−1) stage in the delay elements, (+)output terminal is connected to (−) input terminal of the next stage,and (−) output terminal is connected to (+) input terminal of the nextstage, but for generating oscillation, the (+) output terminal of thelast-stage Cn is connected to (+) input terminal of the first-stage C0,and the (−) output terminal thereof is connected to (−) input terminalof the first-stage C0.

The delay circuit 26 also has (n+1) delay elements C0, C1, . . . Cncascade-arranged in the same manner as the clock generation circuit 16,but the delay elements C0, C1, . . . Cn are connected not in a ringshape but linearly.

The delay elements C0, C1, . . . Cn in the delay circuit 26 have thesame circuit structures as the delay elements C0, C1, . . . Cn of theclock generation circuit 16 so as to being synchronized with the clockperiod in the clock generation circuit 16. That is, the delay elementsof the clock generation circuit 16 and the delay elements of the delaycircuits 26 are the same structures formed on the same semiconductorsubstrate (semiconductor chip). And, the current flowing through each ofthe delay elements C0, C1, . . . Cn in the delay circuit 26 iscontrolled by the same control voltage Vsrc as applied to the clockgeneration circuit 16, and the delay amount is adjusted.

The flipflop 34 operates by detecting the rising edge of a pulse, andtherefore, it is necessary that the delay circuit 26 also alternatelyoutputs “+” and “−” as the two outputs of each of the delay elements C0,C1, . . . Cn so that the rising edge is output.

One example of the circuit structure of each of C0, C1, . . . Cncomposed as the completely differential circuit is shown in FIG. 12.

The p-type transistors Mp1, Mp2 have the common gate and the commonsource potential (power voltage VDD) and connected respectively todrains of n-type MOS transistors Mn1, Mn2. Moreover, the sources of then-type MOS transistors Mn1, Mn2 are connected to the drain of n-type MOStransistor Mn3.

The gate of the transistor Mn1 functions as (+) input terminal, and thegate of the transistor Mn2 functions as (−) input terminal. The drainsof the transistor Mp1 and the transistor Mn1 are connected to each otherto be (−) output terminal. The drains of the transistor Mp2 and thetransistor Mn2 are connected to each other to be (+) output terminal.

By controlling the gate potential Vb1 of the transistors Mp1, Mp2 andthe gate potential Vp2 of the transistor Mn3, the operation current ofthis current can be adjusted, and the delay time can be controlled.

With referring to FIG. 11 again, the (+) terminal output of thefirst-stage delay element C0 and the (+) terminal output of the delayelement C(n+1)/2 are input to the EXOR circuit 11, and the EXOR circuit11 outputs the clock signal clk.

Because one clock period is determined by the delay time of (n+1) delayelements and the delay time of the inverter circuit 70, one clock periodcannot be accurately divided into 1/(n+1) by the influence of the delaytime of the inverter circuit 70, and by contrast, in the structure ofFIG. 11, the oscillation is performed by (n+1) delay elements C0, C1, .. . Cn, and therefore, one period can be accurately divided to 1/(n+1).

Moreover, in the structure of FIG. 11, it is also possible that withoutusing EXOR circuit 11, the clock generation circuit is composed only byC(n+1)/2 delay elements C0, C1, . . . C(n+1)/2 as shown in FIG. 13. Inthe clock generation circuit 17, by decreasing the number of the delayelements to be half in comparison to the clock generation circuit shownin FIG. 11, the occupied area in the semiconductor chip can be reduced.

The application example of the digital PWM circuit of each of theabove-described embodiments includes a digital control power source.Hereinafter, a DC-DC converter will be exemplified as the digitalcontrol power source.

FIG. 14 is a schematic view showing a structure example of the DC-DCconverter, and the DC-DC converter has switching elements Q1, Q2, aninductor L, a condenser C, a control circuit switching ON/OFF of theswitching elements Q1, Q2, and so forth.

The DC-DC converter is a voltage step down DC-DC converter (buckconverter) for obtaining (average) output voltage Vout lower than theinput voltage Vin by alternately switching ON/OFF of the high-sideswitching element Q1 and the low-side switching element Q2.

The elements (a comparator 3, an A/D conversion circuit 4, a PID(proportional-integral-derivative) compensator 5, a digital PWM circuit6, and the switching elements Q1, Q2) surrounded by dashed line in FIG.14 are composed as IC30 that is one chip (or one package).

Each of the switching elements Q1, Q2 is, for example, power MOSFET(Metal-Oxide-Semiconductor Field Effect Transistor), and each of thegate terminals is connected to the digital PWM circuit 6.

The source-drain of the high-side switching element Q1 is connectedbetween the input voltage source 1 and the switch node 40, and thesource-drain of the low-side switching element Q2 is connected betweenthe switch node 40 and the ground. The switch node 40 that is aconnective point of both of the switching elements Q1, Q2 is connectedto a load 2 through the inductor L. Moreover, as a filter element fornot largely changing the output voltage in a short time, a condenser Cis connected between the output side of the inductor L and the ground.

For controlling ON/OFF of the switching elements Q1, Q2, the switchingpulse having an almost inverted phase is provided from the digital PWMcircuit 6 to the gate of each of the switching elements Q1, Q2.

When the high-side switching element Q1 is ON and the low-side switchingelement Q2 is OFF, the current flows from the input voltage source 1through the switching element Q1 to the inductor L, and the inductorcurrent increases, and energy is accumulated in the inductor L. And,when the high-side switching element Q1 is OFF and the low-sideswitching element Q2 is ON, the inductor L releases the accumulatedenergy, and a return current flowing from the ground through theswitching element Q2 to inductor L flows (the inductor current comes todecrease).

The output voltage Vout is controlled so as to converge to a targetoutput voltage (reference voltage Vref). Specifically, the outputvoltage Vout is input to the comparator 3, and the comparator 3 outputsthe comparison result of the output voltage Vout and the referencevoltage Vref to the A/D conversion circuit 4, and the A/D conversioncircuit 4 outputs an error signal e indicating how much the actualoutput voltage disagrees with the target output voltage, as a digitalsignal to the compensator 5. By receiving this error signal e, thecompensator 5 calculates duty of ON/OFF of the switching elements Q1, Q2and output the duty as a duty command value d to the digital PWM circuit6.

Moreover, when both of the switching elements Q1, Q2 are simultaneouslyset to be in the ON state, very large current (pass-through current)comes to flow through both of the switching elements Q1, Q2 to theground. For avoiding this, in setting the duty of ON/OFF of theswitching elements Q1, Q2, the dead time that is a period in which bothof the switching elements Q1, Q2 are OFF is set.

Hereinafter, with reference to FIGS. 15 to 20 and 38 to 40, specificexamples of the digital PWM circuit will be explained. Each of thedigital PWM circuits 81 to 87 shown as follows corresponds to thedigital PWM circuit 6 shown in FIG. 14.

Fifth Embodiment

FIG. 15 is a schematic view illustrating a structure of a digital PWMcircuit 81 according to a fifth embodiment of the invention.

The digital PWM circuit 81 generates gate control signals according tothe duty command value d transmitted from the compensator 5. The gatecontrol signals Out_m×2 and dl1_clk operate flipflop 71 and generategate signals for switching ON/OFF of the high-side switching element Q1.The gate control signals Out_mx3 and Out_mx4 operate the flipflop 72 andgenerate gate signals for switching ON/OFF of the switching element Q2.

Moreover, the digital PWM circuit 81 further has a clock generationcircuit, which is not shown. This clock generation circuit has a ringoscillator structure in which a plurality of delay elements arecascaded-arranged in a ring shape in the same manner as above-describedembodiments, and the digital PWM circuit 81 operates with beingsynchronized with the clock signal (clock) generated by the clockgeneration circuit. Moreover, the delay elements in the clock generationcircuit and the delay elements D of each of the delay circuits 61 to 63shown in FIG. 15 are formed to have the same structures on the samesemiconductor substrate.

FIG. 16 is a timing chart of main signals in the digital PWM circuit 81.

The counter 46 counts clock (FIG. 16( a)) generated by the clockgeneration circuit one by one (period by period) as shown in FIG. 16(b), and supplies the clock to each of the comparators 48, 49, 57, 58,for example, as the 3-bit count values.

Upper several bits (MSB) out of the duty command value (digital signal)d calculated in the compensator 5 shown in FIG. 14 are supplied to thecomparators 48, 57, 58 through the input resistor 47. Lower several bits(LSB) out of the duty command value d are supplied through the inputresistor 47 to multiplexers MUXA, MUXB, and MUXC that are selectioncircuits for plural inputs and one output.

The comparator 48 outputs the signal dl2_clk (FIG. 16( d)) having apulse width of one clock period to the delay circuit 61 at the timingthat the count value agrees with MSB. This signal dl2_clk transmitsthrough each of the delay elements D sequentially from the first stage,and delay is generated in the rising timing of the output signal of eachof the delay elements D.

The output signal of each of the delay elements D is input to themultiplexer MUXA. The multiplexer MUXA selects one (FIG. 16( e)) ofoutput signals of the respective delay elements on the basis of thesignal LSB and outputs the one to the reset terminal R of the flipflop71.

The comparator 49 outputs the signal dl1_clk (FIG. 16( c)) having apulse width of one clock period to the set terminal S of the flipflop 71at the timing that the count value agree with, for example, “000”.

The flipflop 71 outputs the pulse signal shown in FIG. 16( f) from theoutput terminal Q. This output signal switches from the low level to thehigh level at the timing that the signal dl1_clk (FIG. 16( c) input tothe set terminal S switches from the low level to the high level, andthe high level is held while the output signal Out_mx2 (FIG. 16( e)) ofthe multiplexer MUXA input to the reset terminal R is in the low level,and the high level switches to the low level at the timing that thesignal Out_mx2 input to the reset terminal R switches from the low levelto the high level. The output signal (FIG. 16( f)) of the flipflop 71 issupplied to control terminal (gate) of the high-side switching elementQ1, and based thereon, ON/OFF of the switching element Q1 is switched.

That is, the rising timing of the switching pulse for switching ON/OFFof the high-side switching element Q1 is set at the timing of beingsynchronized with clock signal (clock), and the falling timing is set bythe smaller time resolution than that of the clock signal (clock) byutilizing a signal delay in the delay circuit 61.

For the switching pulse for switching ON/OFF of the low-side switchingelement Q2, both of rising and falling timings are set by smaller timeresolution than that of the clock signal (clock) by using the delaycircuits 62, 63 in order to enhance precision of the above-describeddead time adjustment.

That is, the comparator 57 outputs the signal dl3_clk (FIG. 16( i))having a pulse width of one clock period to the delay circuit 62 at thetiming that the count value agrees with MSB. This signal dl3_clktransmits through each of the delay elements D sequentially from thefirst stage, and delay is generated in the rising timing of the outputsignal of each of the delay elements D.

The output signal of each of the delay elements D in the delay circuit62 is input to the multiplexer MUXB. The multiplexer MUXB selects one(FIG. 16( j)) of output signals of the respective delay elements on thebasis of the signal LSB and outputs the one to the reset terminal R ofthe flipflop 72.

The comparator 58 outputs the signal dl4_clk (FIG. 16( g)) having apulse width of one clock period to the delay circuit 63 at the timingthat the count value agrees with MSB. This signal dl4_clk transmitsthrough each of the delay elements D sequentially from the first stage,and delay is generated in the rising timing of the output signal of eachof the delay elements D.

The output signal of each of the delay elements D in the delay circuit63 is input to the multiplexer MUXC. The multiplexer MUXC selects one(FIG. 16( h)) of output signals of the respective delay elements D onthe basis of the signal LSB and outputs the one to the set terminal S ofthe flipflop 72.

The flipflop 72 outputs the pulse signal shown in FIG. 16( k) from theoutput terminal Q. This output signal switches from the low level to thehigh level at the timing that the output signal out_mx4 (FIG. 16( h) ofthe multiplexer MUXC input to the set terminal S switches from the lowlevel to the high level, and the high level is held while the outputsignal Out_mx3 (FIG. 16( j)) of the multiplexer MUXB input to the resetterminal R is in the low level, and the high level switches to the lowlevel at the timing that the signal Out_mx3 input to the reset terminalR switches from the low level to the high level. The output signal (FIG.16( k)) of the flipflop 72 is supplied to control terminal (gate) of thelow-side switching element Q2, and based thereon, ON/OFF of theswitching element Q2 is switched.

Sixth Embodiment

FIG. 17 is a schematic view illustrating a structure of a digital PWMcircuit 82 according to a sixth embodiment of the invention. The samesigns are appended to the same components as the components shown inFIG. 15.

In this embodiment, the functions of the clock generation circuit andthree delay circuits 61 to 63 in the above-described circuit shown inFIG. 15 is consolidated to one. That is, the delay circuit 64 also has afunction of generating the clock signal (clock).

The delay circuit 64 has a plurality of cascade-arranged delay elementsD and an inverter circuit 65, and has a ring oscillator structure inwhich the output of the last-stage delay element D is input to theinverter circuit 65 and the output of the inverter 65 is input to thefirst-stage delay element D.

Moreover, in this embodiment, an AND gate 66 having the output ofcomparator 48 and the output of the multiplexer MUXA as two inputs, anAND gate 67 having the output of comparator 58 and the output of themultiplexer MUXB as two inputs, and an AND gate 68 having the output ofcomparator 57 and the output of the multiplexer MUXC as two inputs, areprovided. The output of the AND gate 66 is input to the reset terminal Rof the flipflop 71, and the output of the AND gate 67 is input to thereset terminal R of the flipflop 72, and the output of the AND gate 68is input to the set terminal S of the flip-flop 72.

The functions as the entire digital PWM circuit are the same as thecircuit shown in FIG. 15, but in this embodiment, when the count valueinput to the comparator 48 agrees with MSB, the output of thecorresponding multiplexer MUXA is set to be effective, and when thecount value input to the comparator 58 agrees with MSB, the output ofthe corresponding multiplexer MUXB is set to be effective, and when thecount value input to the comparator 57 agrees with MSB, the output ofthe corresponding multiplexer MUXC is set to be effective.

That is, at the timing that both of two inputs of AND gate 66 become inthe high level, the reset signal is transmitted to the reset terminal Rof the flipflop 71, and the output of the flipflop 71 switches from thehigh level to the low level. Moreover, at the timing that both of twoinputs of AND gate 68 become in the high level, the set signal istransmitted to the set terminal S of the flipflop 72, and the output ofthe flipflop 72 rises from the low level to the high level, and at thetiming that both of two inputs of AND gate 67 become in the high level,the reset signal is transmitted to the reset terminal R of the flip-flop72, and the output of the flipflop 72 switches from the high level tothe low level.

As described above, also in this embodiment, the rising timing of theswitching pulse for switching ON/OFF of the high-side switching elementQ1 is set at the timing of being synchronized with the clock signal(clock), and the falling timing thereof is set by the smaller timeresolution than that of the clock signal (clock), and for the switchingpulse for switching ON/OFF of the low-side switching element Q2, both ofthe rising and falling timings are set by the smaller time resolutionthan that of the clock signal (clock).

For the above-described delay circuits (delay lines) 61, 62, 63 in theembodiment shown in FIG. 15, it is required that each of the entiredelay amounts thereof is equal to one period of the clock signal (clock)and that the delay amounts of the respective delay elements D are equal.Currently, it is difficult to satisfy this, and in the structure of FIG.15, the area occupied in one chip tends to be large.

By contrast, in the embodiment shown in FIG. 17, the functionscorresponding to the clock generator and the delay circuits 61 to 63 areconsolidated to one circuit 64, and thereby, the entire delay amount canbe precisely set to be one clock period, and the semiconductor devicecan be composed by the small number of elements, and therefore,variation of the delay amounts of the respective delay elements is alsosuppressed, and the occupied area of the delay elements is also reduced.

That is, in this embodiment, by simplifying the structure, theimprovement of the characteristics and reduction of the area can beachieved.

Seventh Embodiment

FIG. 18 is a schematic view illustrating a structure of a digital PWMcircuit 83 according to a seventh embodiment of the invention. The samesigns are appended to the same components as the components shown inFIG. 17.

In this embodiment, the delay circuit 64 is incorporated as onecomponent of DLL (Delay Locked Loop) or PLL (Phase Locked Loop). In thisDLL/PLL 75, the PD (Phase Detector) 74 detects the phase differencebetween the external clock and the internal clock (clock) generated bythe delay circuit 64 and feedbacks the phase difference to the delaycircuit 64 and synchronizes the internal clock (clock) with the externalclock, and thereby, the phase (frequency) of the internal clock (clock)can be maintained to be constant without depending on change oftreatment, voltage, temperature, or the like. The system of the DC-DCconverter is operated with being synchronized with the internal clock(clock), and thereby, characteristic improvement of the entire systemcan be achieved.

Eighth Embodiment

FIG. 19 is a schematic view illustrating a structure of a digital PWMcircuit 84 according to an eighth embodiment of the invention. The samesigns are appended to the same components as the above-describedembodiment. Moreover, FIG. 37 is a timing chart of main signals in thedigital PWM circuit 84.

In the same manner as the above-described embodiments, to the resetterminal R of the flipflop 71 outputting the high-side switching pulse,an output signal Out_mx2 (FIG. 37( b)) of the multiplexer MUXA is input,and to the set terminal S thereof, the output signal dl1_clk (FIG. 37(a)) of the comparator 49 is input.

To the set terminal S of the flipflop 72 outputting a low-side switchingpulse, the signal (FIG. 37( e)) delaying the signal Out_mx2 by the delayelement 78 is input, and to the reset terminal R thereof, the signal(FIG. 37( f)) delaying the signal dl1_clk by the delay element 77 isinput. The delay amounts of the delay element 77 and the delay element78 can be adjusted.

Moreover, the output signal (FIG. 37( c)) of the high-side flipflop 71is delayed by the delay element 76 (FIG. 37( d)).

In this embodiment, only one delay circuit 61 determining the timeresolution of the digital PWM is used, and the dead time is produced byutilizing signal delay by the delay elements 76 to 78 providedseparately from the delay circuit 61. Specifically, by the differencebetween the delay amount of the delay element 76 and the delay amount ofthe delay element 77, the low-side switching element Q2 becomes OFF, andthe dead time is generated until the high-side switching element Q1becomes ON, and by the difference between the delay amount of the delayelement 76 and the delay amount of the delay element 78, the high-sideswitching element Q1 becomes OFF, and the dead time is generated untilthe low-side switching element Q2 becomes ON. The delay elements 76 to78 are not required to have the same characteristics as the delaycircuit 61, and therefore, the design thereof is easy.

Ninth Embodiment

Like a digital PWM circuit 85 shown in FIG. 20, by using DLL/PLL 75 forthe embodiment shown in FIG. 19, the internal clock (clock) can besynchronized with the external clock and maintained to be constant. Whenthe system of the DC-DC converter is operated with being synchronizedwith the internal clock (clock), the characteristic improvement of theentire system can be achieved.

Tenth Embodiment

FIG. 38 is a circuit diagram illustrating a structure of a digital PWMcircuit 86 according to a tenth embodiment of the invention. Thisdigital PWM circuit 86 has the same structure as the digital PWM circuit81 in the fifth embodiment shown in FIG. 15.

Moreover, FIG. 39 is a timing chart of main signals in the digital PWMcircuit 86 of this embodiment.

D1 of FIG. 39( a) represents a gate driving signal of the high-sideswitching element Q1 in the same manner as above-described FIGS. 16( f),and D2 of FIG. 39( d) represents a gate driving signal of the low-sideswitching element Q2 in the same manner as FIG. 16( k).

The D2_set of FIG. 39( b) represents the output of the multiplexer 102in the FIG. 38, and the D2_set corresponds to the out_mx4 (FIG. 16( h))of the multiplexer MUXC in the circuit of FIG. 15. D2_reset of FIG. 39(c) represents the output of the multiplexer 103 in the circuit of FIG.38, and the D2_reset corresponds to the output Out_mx3 (FIG. 16( j)) ofthe multiplexer MUXB in the circuit of FIG. 15.

Moreover, in FIG. 39, “duty” represents ON time in one period (oneswitching cycle) of D1, and “td1” represents a dead time after D1becomes OFF from ON until D2 becomes ON from OFF, and “td2” represents adead time after D2 becomes OFF from ON until D1 becomes ON from OFF.

The digital PWM circuit 86 according to the tenth embodiment shown inFIG. 38 has three delay circuits 91 to 93 and three multiplexers 101 to103 provided so as to correspond to the delay circuits respectively.

Each of the delay circuits 91 to 93 has a plurality of stages ofcascade-arranged delay elements D. Each of the delay elements D has thesame circuit structure formed on the same semiconductor substrate.Moreover, in the delay circuit 92 and the delay circuit 93, the numbers(stage number) of the delay elements D are the same, and the controlcurrent Isrc2 supplied to each of the delay elements D is also the same.

In the former stages of the respective delay circuits 91 to 93,comparators 48, 57, 58 are provided, and to each of the comparators 48,57, 58, the count value cnt of the counter 46 and, for example, adigital signal D [MSB] with top 3 bits of a 5-bit digital signal areinput.

The counter 46 counts the clock signal clk (FIG. 16( a)) one by one(period by period) as shown in FIG. 16( b), and supplies the signal toeach of the comparators 48, 57, 58 as, for example, 3-bit count value.

At the timing that the count value agrees with the MSB[duty], thecomparator 48 outputs the signal va (corresponding to dl2_clk of FIG.16( d)) having a pulse width of one clock period, to the delay circuit91. The signal va transmits through each of the delay elements Dsequentially from the first stage and generates delay in the risingtiming of the output signal of each of the delay elements D.

The output signal of each of the delay elements D in the delay circuit91 is input to multiplexer 101. The multiplexer 101 selects one from theoutput signals of the respective delay elements D on the basis of thedigital signal LSB[duty] and output the one to the reset terminal R ofthe flipflop 71. The reset pulse corresponds to Out_mx2 of FIG. 16( e).

Moreover, to the set terminal of the flipflop 71, the set pulse(corresponding to dl1_clk of FIG. 16( c)) is input at the timing ofbeing synchronized with the clock signal.

The flipflop 71 outputs the signal D1 (corresponding to signal of FIG.16( f)) shown in FIG. 39( a) from the output terminal Q. This outputsignal switches from the low level to the high level at the timing thatthe set pulse switches from the low level to the high level, and thehigh level is held while the reset pulse (output of the multiplexer 101)is in the low level, and the high level switches to the low level at thetiming that the reset pulse switches from the low level to the highlevel. The output signal D1 of the flipflop 71 is supplied to controlterminal (gate) of the high-side switching element Q1, and basedthereon, ON/OFF of the switching element Q1 is switched.

That is, the rising timing of the switching pulse for switching ON/OFFof the high-side switching element Q1 is set at the timing of beingsynchronized with clock signal (clock), and the falling timing is set bythe smaller time resolution than that of the clock signal by utilizing asignal delay in the delay circuit 91.

For the switching pulse for switching ON/OFF of the low-side switchingelement Q2, both of rising and falling timings are set by smaller timeresolution than that of the clock signal by using the delay circuits 92,93 in order to enhance precision of the above-described dead timeadjustment.

That is, the comparator 57 outputs the signal vb having a pulse width ofone clock period to the delay circuit 92 at the timing that the countvalue agrees with MSB [duty+td1]. This signal vb transmits through eachof the delay elements D sequentially from the first stage, and delay isgenerated in the rising timing of the output signal of each of the delayelements D.

The output signal of each of the delay elements D in the delay circuit92 is input to the multiplexer 102. The multiplexer 102 selects one ofoutput signals of the respective delay elements D on the basis of thesignal LSB [duty+td1] and outputs the one to the set terminal S of theflipflop 72 as the set pulse D2_set (FIG. 39( b)).

The comparator 58 outputs the signal vc having a pulse width of oneclock period to the delay circuit 93 at the timing that the count valueagrees with MSB [1−td2]. This signal vc transmits through each of thedelay elements D of the delay circuit 93 sequentially from the firststage, and delay is generated in the rising timing of the output signalof each of the delay elements D.

The output signal of each of the delay elements D in the delay circuit93 is input to the multiplexer 103. The multiplexer 103 selects one ofoutput signals of the respective delay elements on the basis of thesignal LSB [1−td2] and outputs the one to the reset terminal R of theflipflop 72 as the reset pulse D2_reset (FIG. 39( c)).

The flipflop 72 outputs the signal D2 shown in FIG. 39( d) from theoutput terminal Q. This output signal D2 switches from the low level tothe high level at the timing that the set pulse D2_set switches from thelow level to the high level, and the high level is held while the resetpulse D2_reset is in the low level, and the high level switches to thelow level at the timing that the reset pulse D2_reset switches from thelow level to the high level. The output signal D2 of the flipflop 72 issupplied to control terminal (gate) of the low-side switching elementQ2, and based thereon, ON/OFF of the switching element Q2 is switched.

For enhancing voltage precision of the output voltage in the DC-DCconverter, the driving signal D1 of the high-side switching element Q1is required for being generated in a high time resolution. Therefore,for the delay elements D composing the delay circuit 91 setting thefalling timing of D1, it is required to supply the control current Isrc1that is large to some extent in order to shorten the delay time thereof.

Moreover, for shortening the time resolution with respect to the clockperiod, the number of the delay elements D becomes larger, and whenthere are three such delay circuits, the occupied area of the digitalPWM circuit in the chip becomes large.

On the other hand, in generating the driving signal D2 of the low-sideswitching element Q2, the time resolution being as high as that of thehigh-side driving signal D1 is not required. For example, if the timeresolution of the high-side driving D1 is 100 picoseconds, the timeresolution of the low-side driving signal D2 is sufficient to be about 1nanosecond, which is ten-times lower. This is because even if the ON/OFFtiming of the low-side switching element Q2 is changed by a unit of 100picoseconds, conversion efficiency of the power is not affected.

Accordingly, in this embodiment, when the time resolution of the drivingsignal D1 is dt1 as shown in FIG. 39, the time resolution dt2 of each ofthe D2_set signal determining the rising timing of the driving signal D2and the D2_reset signal determining the falling timing thereof is set tobe longer than dt1 (dt1<dt2).

Specifically, in the circuit of FIG. 38, the current Isrc2 supplied toeach of delay elements D in delay circuits 92, 93 is decreased to besmaller than the current Isrc1 supplied to each of delay elements D ofthe delay circuit 91 (Isrc1>Isrc2). For example, when n is an integer of1 or more, the relation of Isrc1=Isrc2×2^(n) can be provided. However,in this case, the relation that the delay time of each of delay elementsD is proportional to the current supplied to each of the delay elementsD is established. As the current supplied to the delay elements issmaller, the time for charging the gate capacity is required more by theamount thereof, and the delay time per delay element is longer, and thedt1<dt2 can be realized. Moreover, to reduce the current supplied to thedelay circuits 92, 93 is to be capable of making the consumption currentsmaller.

As described above, the delay time becomes longer as the current flowedthrough the delay elements is set to be smaller, but when the delay timebecomes long, there is danger that the total delay amount by the entiredelay elements comes not to be contained in one clock period, andtherefore, in this embodiment, the numbers (stage numbers) of the delayelements in the delay circuits 92, 93 is decreased to be smaller thanthe number (stage number) of the delay elements of the delay circuit 91.That is, when the number of the delay elements of the delay circuit 91is (k+1) and the number of delay elements in each of the delay circuits92, 93 is (m+1), k>M is established. Furthermore, when n is an integerof 1 or more, the relation of (k+1)=(m+1)×2^(n) can be provided. Byreducing the number of the delay elements, not only the occupied area ofthe delay elements can be reduced but also the circuit scale of themultiplexers 102, 103 receiving the outputs of the respective delayelements can be smaller.

That is, according to this embodiment, without affecting the powerconversion efficiency very much, cost down by reduction of powerconsumption or reducing of circuit scale can be achieved.

In the above explanation, in the delay circuit 92 and the delay circuit93, the number of the delay elements is set to be the same m, and thesupplied current is set to be the same Isrc2, but by the two delaycircuits 92, 93, the number of the elements or the supplied current maybe set to be different.

In FIG. 36, in the regenerative period in which the load current iL issmall and the high-side switching element Q1 is OFF and the low-sideswitching element Q2 is ON, there is a mode that the magnetic energy ofthe inductor L becomes 0 at a certain time and that the current flowsfrom the output terminal through the low-side switching element Q2 tothe ground. In this case, not only the charge is lost from the condenserin the output terminal but also conduction loss is caused by ONresistance of Q2 by the current flowing through the switching elementQ2. Thereby, the conversion efficiency of the power is significantlydegraded.

Accordingly, it is desirable that in the period that the current flowsfrom the output terminal through the low-side switching element Q2 tothe ground, Q2 is set to be OFF. By setting Q2 to be OFF, the loss isnot caused because the path of the current does not exist, theconversion efficiency of the power can be improved.

It can be easily thought that for the timing of setting Q2 to be OFF,sufficient time before the magnetic energy of the inductor L becomes 0is taken for setting OFF. In this case, because the magnetic energy ofthe inductor L is not 0, the current continues to flow through theinternal diode of Q2. In this case, because the internal diode has highON voltage in comparison to the MOS structural part, the conduction lossbecomes large compared to the case that the MOS structural part is ON.Therefore, the time resolution of td2 is occasionally required more thanthat of td1 for improving the efficiency. In this case, the number ofthe delay elements D of the delay circuit 93 is set to be larger thanthe number of the delay elements D of the delay circuit 92.

At any rate, in the delay circuits 92, 93, it is necessary that thenumber of the delay elements is set to be smaller than that of the delaycircuit 91 and that the supplied current is also set to be smaller thanthat of the delay circuit 91.

Eleventh Embodiment

FIG. 40 is a circuit diagram illustrating a structure of a digital PWMcircuit 87 according to an eleventh embodiment of the invention.

In the above-described tenth embodiment, as shown in FIG. 39, the setpulse D2_set of D2 counts MSB[duty+td1] by the counter 46, and comparesLSB[duty+td1] therewith by the comparator 57, and inputs the signal tothe delay circuit 92. In this structure, for the dead time td1, anegative value can also be set. The negative value of td1 corresponds tothe case that the falling timing of D2 is set to be before the fallingtiming of D1. When the ON time of the low-side switching element isextremely larger than the OFF time of the high-side switching element,td1 is required to be set to be negative, but in general, the high-sideand low-side switching elements whose ON time and OFF time are extremelydifferent are not used. Accordingly, td1 rarely becomes negative.

Accordingly, in this embodiment, as shown in FIG. 40, the signal (outputof the multiplexer 101) generating the reset pulse of D1 is input to thedelay circuit 92 generating the set pulse D2_set of D2. Thereby, risingof D2 is generated on the basis of falling of D2, and therefore, thecounter 46 and the comparator 57 at the former stage of the delaycircuit 92 becomes unnecessary, and the circuit scale can be smallerthan that of the tenth embodiment.

In this embodiment, in the same manner as the tenth embodiment, in eachof the delay circuits 92, 93, the number of the delay elements is set tobe smaller and the supplied current is also set to be smaller than thatof the delay circuit 91, and thereby, the time resolution dt2 of each ofthe D2_set signal determining the rising timing of the driving signal D2and the D2_reset signal determining the falling timing thereof may belonger than the time resolution dt1 of the driving signal D1 (dt1<dt2),and also, dt2=dt1 is possible.

Moreover, in the same manner as the tenth embodiment, in the delaycircuit 92 and the delay circuit 93, the number of the delay elements isset to be the same m, and the supplied current is also set to be thesame Isrc2, but the number of the elements or the supplied current maybe set to be different between the two delay circuits 92, 93.

Twelfth Embodiment

Next, one specific example of the structure corresponding to thecomparator 3 and the A/D conversion circuit 4 in the structure shown inthe above-described FIG. 14 will be explained. FIG. 21 is a schematicview illustrating a structure of DC-DC converter according to a twelfthembodiment of the invention. The same signs are appended to the samecomponents as the above-described embodiments.

The output voltage Vout of the switching power circuit having theswitching elements Q1, Q2 and the inductor L and the condenser C iscompared to the target output voltage in the error signal generationcircuit having the D/A conversion circuit 7 and the comparators 3 a to 3f and the A/D conversion circuit 4, and the error signal e correspondingto the difference between the output voltage Vout and the target outputvoltage is generated.

Specifically, the output voltage Vout is input to the comparator 3 a to3 f and compared with the target output voltage (reference voltageVref). In the D/A conversion circuit 7, as shown in FIG. 22, a pluralityof stages (six in the example shown in the figure) of Bin (thresholds)are set, and supplied to each of the comparators 3 a to 3 f as theanalog voltages.

The comparator 3 a compares Vref and (Vref+q/2), and the comparator 3 bcompares Vref and (Vref−q/2), and the comparator 3 c compares Vref and(Vref+3 q/2), and the comparator 3 d compares Vref and (Vref−3 q/2), andthe comparator 3 e compares Vref and (Vref+5 q/2), and the comparator 3f compares Vref and (Vref−5 q/2).

The A/D conversion circuit 4 receives the comparison result of thecomparators 3 a to 3 f, and outputs the control parameter forcompensating the disagreement amount of the output voltage Vout withrespect to the target output voltage (reference voltage Vref), as theerror signal (digital value) e to the compensator 5.

The A/D conversion circuit 4 determines the error signal e withreference to Look Up Table in which the corresponding relation betweenBin (q/2, −q/2, 3 q/2, −3 q/2, 5 q/2, −5 q/2) and the control parameter(q, −q, 3 q, −3 q, 5 q, −5 q) is preliminarily written.

When the output voltage Vout becomes larger than Vref and disagrees toreach (Vref+q/2), the error signal e corresponding to the controlparameter “q” is output, and when the output voltage Vout becomessmaller than Vref and disagrees to reach (Vref−q/2), the error signal ecorresponding to the control parameter “−q” is output, and when theoutput voltage Vout becomes larger than Vref and disagrees to reach(Vref+3 q/2), the error signal e corresponding to the control parameter“2 q” is output, and when the output voltage Vout becomes smaller thanVref and disagrees to reach (Vref−3 q/2), the error signal ecorresponding to the control parameter “−2 q” is output, and when theoutput voltage Vout becomes larger than Vref and disagrees to reach(Vref+5 q/2), the error signal e corresponding to the control parameter“3 q” is output, and when the output voltage Vout becomes smaller thanVref and disagrees to reach (Vref−5 q/2), the error signal ecorresponding to the control parameter “−3 q” is output.

The compensator 5 calculates the duty command value d on the basis ofthe error signal e and outputs the duty command value d to the digitalPWM circuit 6. For example, with reference to Look Up Table in which thecorresponding relation between the error signal e and the duty commandvalue d calculated with respect to the error signal is preliminarilywritten, the duty command value d is determined.

Like the above-described embodiments, the digital PWM circuit 6generates a switching pulse provided to the gate of the high-sideswitching element Q1 and a switching pulse provided to the gate of thelow side switching element Q2, and supplies the switching pulses to eachof the switching elements Q1, Q2.

Thirteenth Embodiment

Next, with reference to FIGS. 24 to 26, a thirteenth embodiment of theinvention will be explained. FIGS. 24, 25, 26 correspond toabove-described FIGS. 21, 22, 23, respectively.

When the output voltage Vout is changed rapidly by rapid change of thecharge, if the disagreement amount with respect to Vref comes to exceedthe uppermost or lowermost Bin, the control parameter becomes quite thesame (3 q or −3 q) in the region thereof no matter how far Vout isseparate from Vref, and therefore, when Vout disagrees so as to beextremely larger than (Vref+5 q/2) or when Vout disagrees so as to beextremely smaller than (Vref−5 q/2), Vout does not converge to Vref onlyby the control of one time of the control parameter (3 q or −3 q), andthe control by the control parameter (3 q or −3 q) becomes repeated atseveral times, and time is required for making Vout converge to Vref.That is, if Bin width and the control parameter are fixed, the controlresponse with respect to large rapid change of the charge is degraded.

Accordingly, in this embodiment, to the uppermost Bin (Vref+5 q/2) andthe lowermost Bin (Vref−5 q/2), parameters of (+a) and (−a) can beadded, respectively. Furthermore, as shown in FIG. 26, the controlparameter (3 q+b) corresponding to Bin [(Vref+5 q/2)+a] and the controlparameter (−3 q-b) corresponding to Bin [(Vref−5 q/2)−a] are written inLook Up Table. For a, a plurality of values can be set, andcorrespondingly, b can be a plurality of values.

First, for example, a=0 and b=0 are set. When the load rapidly decreasesand the output voltage Vout exceeds (Vref+5 q/2), the value of a is setto be a larger value (correspondingly the value of b becomes larger),the uppermost Bin is changed from (Vref+5 q/2) to [(Vref+5 q/2)+a], andat the same time, the control parameter (3 q+b) corresponding toBin[(Vref+5 q/2)+a] is read from Look Up Table, and the error signal ecorresponding to the control parameter (3 q+b) is output.

That is, when the output voltage Vout disagrees to be larger than Vref,by enhancing Bin width correspondingly, the larger control parameter canbe used and the output voltage Vout can be made to converge to Vref in ashort time, and response in the control operation of setting the outputvoltage Vout to be a desired target value can be enhanced.

Also, the case that the load rapidly increases can be thought similarly,and when the output voltage Vout becomes smaller than (Vref−5 q/2), thevalue of a (absolute value) is set to be larger than 0, and thelowermost Bin is changed from (Vref−5 q/2) to [(Vref−5 q/2)−a], andsimultaneously, the control parameter (−3 q−b) corresponding to Bin[(Vref−5 q/2)−a] is read from Look Up Table, and the error signal ecorresponding to control parameter (−3 q−b) is output.

As described above, according to this embodiment, in operating the powercircuit, the uppermost or lowermost Bin width is changed according tothe change width of the output voltage Vout, and the control parameteris changed to be a larger control parameter, and thereby, the responsewith respect to the large rapid change of the charge (convergentproperty of Vout to the target value) can be improved.

In the above-described embodiment, the specific examples of changinguppermost and lowermost Bin widths has been exemplified, but byappropriately changing the Bin widths of the other positions similarly,the improvement of response can be expected.

Next, the embodiment that a plurality of DC-DC converters are connectedin parallel and driven in parallel will be explained.

Fourteenth Embodiment

FIG. 27 is a schematic view showing a structure of a digital controlpower source according to a fourteenth embodiment.

A plurality of DC-DC converters 50-1, 50-2, . . . 50-N are connected inparallel between the input voltage source 1 (see, FIG. 14) and theoutput terminal. The output side of the inductor L of each of the DC-DCconverters 50-1, 50-2, . . . 50-N is connected in parallel to the commonoutput line 90. In the output line 90, a condenser C and the load 2 arecommonly connected to each of the DC-DC converters 50-1, 50-2, . . .50-N.

The power ICs 30-1, 30-2, . . . 30-N of the respective DC-DC converters50-1, 50-2, . . . 50-N correspond to the power IC 30 surrounded bydashed line in FIG. 14. As the digital PWM circuit 6 and the comparator3 and the A/D conversion circuit 4 in each of the power ICs 30-1, 30-2,. . . 30-N, the ones having the structures described above withreference to FIGS. 1 to 26 and 38 to 40 can be appropriately used.

The power ICs 30-1, 30-2, . . . 30-N have the same structures, and theinductors L also have the same structures, and therefore, the DC-DCconverters 50-1, 50-2, . . . 50-N have the same structures, and the sameoutput voltage can be output in principle. Furthermore, each of theDC-DC converters 50-1, 50-2, . . . 50-N (the power ICs 30-1, 30-2, . . .30-N) has the capability of becoming the master (function of beingcapable of supplying reference signal or the like to the otherconverters).

When the output current of each of the DC-DC converters 50-1, 50-2, . .. 50-N is, for example, 10 amperes, if, for example, 10 converters areconnected in parallel and operated in parallel, a power source having anoutput current of 100 amperes can be composed. This can be moreinsufficient than the case that one DC-DC converter having an outputcurrent of 100 amperes is used.

Moreover, the output voltage of each of the DC-DC converters 50-1, 50-2,. . . 50-N has a ripple as shown in FIG. 28A, but when each of the DC-DCconverters 50-1, 50-2, . . . 50-N is operated (multiphase-operated) sothat phases of output voltages of the respective DC-DC converters 50-1,50-2, . . . 50-N come to disagree little by little as shown in FIG. 28B,the ripple can be reduced.

The power ICs 30-1, 30-2, . . . 30-N are connected to a common (one)data bus 22. Furthermore, the power ICs 30-1, 30-2, . . . 30-N areconnected to a common (one) synchronization signal line 21. Throughthem, the power ICs 30-1, 30-2, . . . 30-N communicate to one another.

First, on start-up, the power IC to be a reference of the output voltagephase 0° is determined. Here, for example, the power IC 30-1 is made tobe the power IC. The power IC outputs the internal reference clock fromSYNC terminal (not shown) to the synchronization signal line 21, and theother power ICs 30-2, . . . 30-N receive the reference clock from eachof the SYNC terminals through the synchronization signal line 21, andoperates with being synchronized with the reference clock by using, forexample, the above-described structure such as PLL. That is, all of theDC-DC converters 50-1, 50-2, . . . 50-N operate with being synchronizedwith the same reference dock.

Moreover, the power IC 30-1 indicates a phase shift value of the outputvoltage through the data bus 22 to the other power ICs 30-2, . . . 30-N.Each of the power ICs 30-2, . . . 30-N except for the power IC 30-1receives the phase shift value through the data bus 22, and as shown inFIG. 28B, phases of output voltages of the respective power ICs 30-1,30-2, . . . 30-N are made to disagree with one another, and the rippleis reduced.

All of the power ICs 30-1, 30-2, . . . 30-N operate with beingsynchronized with the reference clock of the power IC 30-1. Therefore,if a trouble is caused in the reference clock inside the power IC 30-1,the entire system becomes failed. For preventing this, the power IC 30-1has a function of detecting its own trouble, and if the trouble isdetected, the sequence data of which power IC is next made to functionas the master is indicated through the data bus 22. Therefore, all ofthe power ICs 30-1, 30-2, . . . 30-N can be the reference IC having aphase of 0° and have a function of indicating the phase shift value tothe other power ICs. If a trouble is caused in the power IC functioningas the master, another normally operating power IC functions as themaster instead of the troubled power IC, and therefore, the reliabilityof the entire system can be enhanced.

As described above, in this embodiment, the power ICs communicate to oneanother and can autonomously set the interleaving, and therefore, evenif a trouble is caused in any one power IC, the system of the entireDC-DC converters is not failed, and therefore, the reliability can beenhanced.

Fifteenth Embodiment

In a digital control power source in which N DC-DC converters (powerICs) having the same structures are driven in parallel, for maximizingefficiency (output power/input power), how each of the power ICs isoperated is important. In particular, when the entire output currentcomes to decrease in such a case as light load, compared to the case inwhich the current of the entire current I divided by N (I/N) is flowedto each of N power ICs, the efficiency becomes high in the case in whichthe operation of some power ICs out of the N power ICs is stopped andonly Nop (<N) power ICs are operated and the current of (I/Nop) isflowed to each of the operating power ICs.

In general, when the output current becomes small in the DC-DCconverter, the efficiency becomes lower as shown in FIG. 29. In thiscase, consumption current of the control circuit or power for chargingand discharging the gate of the power stage (switching elements Q1, Q2)does not change and does not decrease even when the output current ofthe DC-DC converter changes. Accordingly, as the output power comes todecrease, the efficiency (output power/input power) lowers.

When the entire output current decreases and therewith the outputcurrent of each of N power ICs lowers and the efficiency of the currentbecomes lower than the peak, operation of at least one or more power ICsout of N power ICs is stopped and the current flowing through theindividual operating power source ICs is held to be high, and thereby,the efficiency can be held to be high. The power ICs whose operation isstopped are naturally the power ICs except for the power IC functioningas the master.

In the example of FIG. 29, when the value of (entire current I/operation number Nop) becomes smaller than the current Imax in which theefficiency becomes the peak, if the operation number Nop is reduced tosuppress the current value (I/Nop) flowing through each of the power ICsto be in the vicinity of Imax, the efficiency can be held to be high.

When Nop power ICs are operating now and the entire output current is I,the current of approximately (I/Nop) flows through each of the powerICs. And, when (I/Nop) becomes smaller than the preliminarily determinedcurrent value, X power ICs out of Nop power ICs are stopped. In orderthat the residual (Nop-X) power ICs share and continuously flow theentire current I, the value of the current flowing through each of the(Nop-X) power ICs is set to be [Nop/(Nop−X)]-times larger.

For the power IC whose operation is stopped, because operations of alarge part of the control circuit and the power stage (switchingelements Q1, Q2) are stopped, the power consumption becomes almost zero.The operating circuit comes to be only the part waiting so as to becapable of operating by receiving the input of the signal when thecurrent increases.

In determining the number Nop of the power ICs made to operate, asdescribe later, (I/Nop) by which each of the power ICs maintains theoperation by the continuous mode is preferable.

In general, a DC-DC converter has operation regions of a continuous modeand discontinuous mode. The continuous mode is an operation mode inwhich the inductor current flowing to the direction of supplying thecurrent to the load is larger than 0, and the discontinuous mode is anoperation mode including a period in which the inductor current flowingto the direction of supplying the current to the load becomes 0.

In particular, if the current becomes small in the case of light load,when the high-side switching element Q1 is OFF and the low-sideswitching element Q2 is ON, the inductor current becomes zero ornegative current in which the current is not supplied to the load, andthe operation becomes the discontinuous mode. In the example shown inFIG. 29, when the current (I/Nop) flowing through each of Nop operatingpower ICs (each of DC-DC converters) becomes I₀ or less, the operationbecomes the discontinuous mode.

As shown in FIG. 29, in the region that (I/Nop) is I₀ or less, theefficiency lowers more and more, and in the region of larger than I₀,there is the value Imax of (I/Nop) of maximizing the efficiency.Accordingly, by adjusting the number Nop of the operating power ICs sothat each of the power ICs always operate in the continuous mode (sothat I/Nop becomes larger than I₀), lowering of the efficiency issuppressed and the operation in the vicinity of the maximum efficiencybecomes possible.

Moreover, in particular, in the digital control DC-DC converter, thetransfer function in the control system is different between thecontinuous mode and the discontinuous mode, and therefore, the controlis required to be changed, but by setting the operating power ICs to bein a state of the continuous mode, the same transfer function can beused for the control system of each of the power ICs, and the controlbecomes simple.

Sixteenth Embodiment

In the DC-DC converters made to operate in parallel, it is desirablethat as described above, the output current of each of the DC-DCconverters becomes (I/ operation number Nop) when the current flowingthrough the load is I, but by manufacturing variation or the likedespite the power ICs the same structures, there are variations of theinternal reference voltage (reference voltage Vref and the offsetvoltage of the A/D conversion circuit 4, and also, elementcharacteristics such as each of the output FETs (switching elements Q1,Q2) or the LC filter vary, and therefore, unless the output voltagecontrol loop is set to be one and positive current balance control isperformed, it is difficult to flow the uniform current through each ofthe DC-DC converters (each of power ICs).

When each of the power ICs regulates the output (controls the output tothe target voltage) by the control loop using its own output voltagecontrol circuit (the comparator 3, the A/D conversion circuit 4, thecompensator 5, and so forth), for example only Vref of the power IC 30-1varies to be higher with respect to the other power ICs 30-2, . . .30-N, almost all of the entire current I comes to concentrate on theDC-DC converter 50-1, and the current flowing through the other DC-DCconverters 50-2, . . . 50-N becomes almost zero.

FIG. 30 is a schematic view showing a structure of a digital controlpower source according to a sixteenth embodiment of the invention. Thesame signs are appended to the same components as above-described FIG.27, and the detailed explanation thereof will be omitted.

Each of the power ICs 30-1, 30-2, . . . 30-N are connected in parallelto a common (one) error share bus 23. Each of the operating power ICscontrols ON/OFF duty of the switching elements Q1, Q2 on the basis ofthe error signal e obtained in the error signal generation circuit (thecomparator 3, the A/D conversion circuit 4) as explained in theabove-described embodiment. In this embodiment, the error signal eobtained in any one of the power ICs is shared and used by the operatingentire power IC.

For example, if the power IC 30-1 is made to be the power IC functioningas the master, the power IC 30-1 calculates the duty command value d bythe compensator 5 by using the error signal e obtained by its own A/Dconversion circuit 4 and controls the output voltage on the basis of theduty command value d, and transmits the error signal e through the errorshare bus 23 to the other operating power ICs. The other power ICscalculate the duty command value d by the respective compensators 5 onthe basis of the error signal e, and control ON/OFF of the switchingelements Q1, Q2, and thereby control the output current.

That is, because ON/OFF control of the switching elements Q1, Q2 isperformed based on the common error signal e in each of the operatingpower ICs, independent from the characteristic variation of the errorsignal generation circuit part (the comparator 3, the A/D conversioncircuit 4, and so forth) among the power ICs, the output current can beevenly distributed among the power ICs, and the current can be preventedfrom concentrating on one Power IC.

Not the error signal e but the duty command value d may be shared andused among the operating entire power ICs. That is, the power IC 30-1calculates the duty command value d by the compensator 5 by using theerror signal e obtained by its own A/D conversion circuit 4 and controlthe output voltage and transmits the duty command value d through theerror share bus 23 to the other operating power ICs. The other power ICscontrol On/OFF of the respective switching elements Q1, Q2 on the basisof the duty command value d.

Because ON/OFF control of the switching elements Q1, Q2 is performedbased on the common duty command value d in each of the operating powerICs, independent from the characteristic variation of the comparator 3and the A/D conversion circuit 4 and the compensator 5 and so forthamong the power ICs, the output current can be evenly distributed amongthe power ICs, and the current can be prevented from concentrating onone Power IC.

In performing the above-described control, it is necessary to considerthe delay time in which the power IC 30-1 functioning as the mastercalculates the voltage compensation information (the error signal e orthe duty command vale d) and then transmits the information to the otheroperating power IC and then the other power ICs drive ON/OFF of theswitching elements Q1, Q2 on the basis of the received voltagecompensation information.

Here, in the multiphase operation that each of the power ICs makes theoutput phases disagree with one another, the power IC 30-1 functioningas the master calculates the above-described voltage compensationinformation at the times of the number of the operating power ICs (atNop times) and transmits the voltage compensation informationsequentially from smaller phases with respect to the phase 0° of thepower IC 30-1, and thereby, the delay time between the timing that thepower IC 30-1 transmits the voltage compensation information to thetarget power IC and the timing that the switching elements Q1, Q2 of thepower ICs receiving the information start to switch is suppressed to besmall, and thereby, the control response to the target output voltagecan be enhanced. And, additionally, by the multiphase operation, powersystem in which the output ripple is suppressed can be realized.

Moreover, in the transient response, by using not the voltagecompensation information from the power IC 30-1 but its own comparator 3and the A/D conversion circuit 4 and the compensator 5, the error signale or the duty command value d is calculated to perform the transientresponse, and thereby, the response property can be more improved. Inthis case, in the static state, in the power ICs 30-2, . . . 30-N exceptfor the power IC 30-1, it is preferable to adjust the internal referencevoltage (reference voltage Vref) so that the error signal e that is thesame as the error signal e transmitted from the power IC 30-1 can begenerated. Thereby, because the variation of the reference voltage ofeach of the power IC can be compensated, the error among the phasecurrents can be smaller.

Seventeenth Embodiment

By the above-described sixteenth embodiment, the output current of eachof the power ICs is balanced uniformly to some extent, but thecharacteristic variation of the output FETs (switching elements Q1, Q2)or the LC filter of each of the power ICs cannot be compensated, andtherefore, higher precise current balance control becomes required.

FIG. 31 is a schematic view showing a structure of a digital controlpower source according to a seventeenth embodiment of the invention. Thesame signs are appended to the same components as the FIGS. 27 30, andthe detailed explanation thereof will be omitted.

In this embodiment, in addition of the above-described synchronizationsignal line 21 and the error share bus 23, to a common (one) currentshare bus 24, the power ICs 30-1, 30-2, . . . 30-N are connected inparallel. Each of the power ICs 30-1, 30-2, . . . 30-N transmits andreceives the current value data through the current share bus 24 withbeing synchronized with the clock signal supplied through thesynchronization signal line 21.

The power ICs 30-1, 30-2, . . . , 30-N have a driver Dv and a receiverRv connected to the current share bus 24 as shown in FIG. 32. In FIG.32, each of the power ICs 30-1, 30-2, . . . 30-N is shownrepresentatively by the sign 30.

To the driver Dv, the signal T of its own current value converted into apulse width is input. FIG. 33 shows an example that the current value isconverted into the high-level pulse width W1 with being synchronizedwith the clock signal X. The signal T is inverted in the driver Dv andoutput to the current share bus 24, and also input to its own receiverRv.

The driver is a so-called an open drain type. In this embodiment, theopen drain output terminal of the driver Dv of each of the power ICs isconnected in parallel to the current share bus 24, and additionally,pulled up to the power source by the resistance R shown in FIG. 31, andthe function of the negative-logic wired OR (wired OR) is realized. Thatis, when the driver Dv of any one of the power ICs is outputting “Low”,the current share bus 24 becomes in the “Low” level.

This will be explained by using FIG. 34 as the example. Each of thepower ICs is synchronized with the signal synchronized with theswitching cycle and outputs the pulse signals Y-1, Y-2, . . . Y-n havingpulse widths corresponding to the respective current values to thecurrent share bus 24. When the current value is converted to thehigh-level pulse width W1 like the signal T shown in FIG. 33, thecurrent value is inverted by the driver Dv and therefore the low-levelwidth corresponds to the current value in each of the signals Y-1, Y-2,. . . Y-n.

In the example shown in FIG. 34, because the low-level pulse width ofthe signal Y-2 is the longest, the signal Y in the current share bus 24comes to have the same wave shape as the signal Y-2 by the negativelogic wired OR, and the common signal Y is input to the receiver Rv ofeach of the power ICs.

The signal Y is converted in the receiver Rv to be the signal S, andfrom the high-level pulse width of this signal 5, the maximum currentvalue can be detected in the output current values of the respectiveoperating power ICs. That is, each of the operating power ICs can sharethe maximum current value data as the share current value data.

The minimum current value of the output current values of the operatingpower ICs may be shared among the power ICs as the share current valuedata.

That is, when each of the operating power ICs converts its own currentvalue into the low-level pulse width W2 as shown in FIG. 35, the signalT is inverted by the driver Dv and therefore the high-level widthcorresponds to the current value in the each of the signals Y-1, Y-2, .. . Y-n in FIG. 34.

And, the signal Y comes to have the same wave shape as the signal havingthe largest low-level pulse width, and conversely, the signal having thesmallest high-level pulse width is shared among the power ICs as thesignal Y. In this case, because the current value is converted to thehigh-level width, it corresponds to the minimum current value that thehigh-level width is the smallest.

In this embodiment, for example, the power ICs are connected in a ringshape to compose a token ring, and the transmission sequence of thecurrent value data is determined on start-up, the power ICs taking inthe token (transmission right) are made to sequentially transmit thecurrent value data to the current share bus 24.

The time T required for recognizing the current value data of the entireoperating phases can be represented by T=Nphase X Ts/ Xphase, in whichNphase is the number of operation phases and Ts is switching period andXphase is the number of phases of the current value data shared in oneswitching cycle.

By using the share current value data obtained as described above, eachthe operating power ICs composes the current compensation loop as shownin FIG. 36 in addition of the above-described voltage compensation loop,and compensates the duty command value d obtained by the voltagecompensation loop.

That is, each of the power ICs compares the share current value dataobtained through current share bus 24 and the own output current value(inductor current value) iL by the comparator 12, and the comparisonresult is output to the current compensator 13, and the currentcompensator 13 calculates the duty compensation value Δd on the basis ofthe comparison result. The Δd corresponds to the duty control amount forcontrolling iL to be the current corresponding to the above-describedshare current value data.

When the current error that each of the power ICs subtracts its owncurrent value from the share current value is ei, the transfer functionGdi(z) from ei to Ad can be exemplified in the following formulas 1-1,1-2. In this case, b0 to b2 are constants.

$\begin{matrix}{{G_{di}(z)} = \frac{b_{0}z}{z - 1}} & {1\text{-}1} \\{{G_{di}(z)} = \frac{{b_{0}z^{2}} + {b_{1}z} + b_{2}}{z\left( {z - 1} \right)}} & {1\text{-}2}\end{matrix}$

When the transfer function Gdi(z) is known, by an inverse ztransformation of transforming the transfer function (frequency axis) toa difference equation, the output (Δd) can be calculated from the input(ei). In the case of Gdi(z) of the above-described formulas 1-1, 1-2, Adbecomes the following formulas 2-1, 2-2, respectively. [n] representsthe number n of the sample data. In the above-described treatment, afterthe A/D conversion treatment, the signal is discretized.

Δd[n]≦Δd[n−1]+b ₀ e[n]  2-1

Δd[n]=Δd[n−1]+b ₀ e[n]+b ₁ e[n−1]+b ₂ e[n−2]  2-2

Δd calculated by the current compensator 13 is biased in the accumulator18 with respect to the duty command value d calculated in the voltagecompensator 5, and (d+Δd) is supplied to the digital PWM circuit 6 asthe duty command value, if only the current compensation loop is viewed,the control is that the current is uniformed to the maximum currentvalue or the minimum current value among the power ICs, but each of thepower ICs individually performs the duty control so that the voltagebecomes the target voltage (reference voltage Vref) by theabove-described voltage compensation loop, and therefore, by performingthe switching control of the switching elements Q1, Q2 on the basis of(d+Δd), the output current of each of the power ICs is compensated tothe direction to becoming smaller (larger) when the current is too large(too small), and therewith, the current is uniformed among the powerICs.

That is, each of the power ICs shares the voltage compensationinformation (the error signal e or the duty command value d) through theerror share bus 23, and thereby the current nonuniformity due tocharacteristic variation of the compensator 3 and the A/D conversioncircuit 4 and the voltage compensator 5 and so forth can be compensated,and additionally, the share current value data is shared through thecurrent share bus 24, and thereby, the current nonuniformity due tocharacteristic variation of the output FETs (switching elements Q1, Q2)or LC filter can be compensated. As a result, the current balancecontrol among the power ICs can be performed higher-precisely and thecurrent concentration on one power IC can be avoided.

It is not necessary that the current share bus 24 and the error sharebus 23 are separated as the hardware, but a common bus can also be usedby sectioning the time so that the voltage compensation information iscommunicated in one time period of the inside of the switching cycle andthat the share current value data is communicated in the other timeperiod.

In this embodiment, in sharing the share current value data required forthe control to evenly uniform the output current among a plurality ofDC-DC converters (power ICs) connected in parallel, high-speed digitalbus is not used, and the power consumption is small. Moreover, thereference voltage Vref is not shared among the power ICs, and theincrease of the number of the terminals can be suppressed, andfurthermore, all of the communications are performed by the digitalmethod, and therefore, are difficult to be affected by the wiring delay.

The abnormal state (error) of the power system includes inputovervoltage, input low-voltage, input overcurrent, input overpower,output overvoltage, output low-voltage, output overcurrent, outputoverpower, output tracking error, overtemperature, low-temperature,nonvolatile memory read error, and so forth.

In recent years, market requirement for power management is enhanced,and it is necessary that the control (power-system protection) methodsin the various power system abnormalities can be finely set for each ofthe abnormal states.

On the other hand, for various errors, the power system can be protectedby controlling the output-stage FET. For example, the case that theoutput overcurrent error and the output overvoltage error aresimultaneously caused is thought. When the power IC is set so that theoutput current is maintained to be constant in the output overcurrenterror and so that the output is stopped in the output overvoltage,because the controllable part is only the output-stage FET, theoperation simultaneously satisfying the above-described settings cannotbe realized and contradiction is caused.

Accordingly, for avoiding this, in the embodiments of this invention,the priority sequence is determined for each of errors in each of theabove-described power systems, and when a plurality of errors are causedat the same time, the errors are addressed from higher prioritysequence. The priority sequence can be set in the sequence of higherrisk for the power system, and can also be programmable through acommunication bus or a nonvolatile memory.

As described above, the embodiments of this invention have beenexplained with reference to the specific examples. However, thisinvention is not limited thereto, and various modifications based on thetechnical ideas of this invention are possible.

In the above-described embodiments, the voltage step down DC-DCconverter has been exemplified as the digital control power, but thisinvention is not limited thereto and is applicable to voltage step upDC-DC converter or other voltage conversion circuits. Moreover, thedigital PWM circuit of this invention illustrated in FIGS. 1 to 20 isapplicable to other applications such as motor driver and LED (LightEmitting Diode).

According to a fourth aspect of the invention, there is provided asemiconductor device comprising: a switching power circuit having aswitching element; an error signal generation circuit for detecting anoutput voltage of the switching power circuit and comparing the outputvoltage with a target voltage and generating a error signal on the basisof the comparison result thereof; and a digital PWM circuit forsupplying a pulse signal whose duty is determined based on the errorsignal to compensate disagreement of the output voltage with respect tothe target voltage, to a control terminal of the switching element, theerror signal generation circuit outputting a control parameterdetermined based on a corresponding relation between a threshold set tohave a plurality of stages according to disagreement width with respectto the target voltage and the control parameter set with according tothe threshold, a width of the threshold being changed in operating theswitching power circuit, and the control parameter changing withcorresponding to the change of the width of the threshold.

There is provided a device according to the fourth aspect, wherein whenthe output voltage becomes larger than the uppermost threshold, thewidth of the uppermost threshold is changed to be larger.

There is provided a device according to the fourth aspect, wherein whenthe output voltage becomes smaller than the lowermost threshold, thewidth of the lowermost threshold is changed to be smaller.

There is provided a device according to the fourth aspect, furthercomprising a compensator calculating the duty on the basis of the errorsignal.

According to a fifth aspect of the invention, there is provided asemiconductor device comprising a plurality of digital control powersources having same structures connected in parallel with respect to acommon output line, a current flowing each of the operating digitalcontrol power sources out of the plurality of digital control powersources being controlled so that each of the operating digital controlpower sources maintains operation in a continuous mode.

According to a sixth aspect of the invention, there is provided asemiconductor device comprising: a plurality of digital control powersources having same structures connected in parallel with respect to acommon output line; and a signal line connected in parallel so that theplurality of digital control power source can communicate with oneanother, each of the digital control power sources having an outputvoltage control circuit controlling each of output voltages to be atarget voltage, voltage compensation information obtained by the outputvoltage control circuit of any one of the operating digital controlpower sources out of the plurality of digital control power sourcesbeing shared among all of the operating digital control power sourcesthrough the signal line, the operating digital control power sourcescontrolling the output voltage on the basis of the shared voltagecompensation information.

There is provided a device according to the sixth aspect, wherein theoutput voltage control circuit has an error signal generation circuitfor detecting the output voltage and comparing the output voltage withthe target voltage and generating an error signal corresponding to thedisagree amount of the output voltage with respect to the targetvoltage, and each of the digital control power sources sharing the errorsignal.

There is provided a device according to the sixth aspect, wherein eachof the digital control power sources has a switching element, and theoutput voltage control circuit has: an error signal generation circuitfor detecting the output voltage and comparing the output voltage withthe target voltage and generating an error signal corresponding to thedisagree amount of the output voltage with respect to the targetvoltage; and a compensator for calculating duty of pulse signalswitching ON/OFF of the switching element on the basis of the errorsignal, and each of the digital control power sources sharing the duty.

According to a seventh aspect of the invention, there is provided asemiconductor device comprising: a plurality of digital control powersources having same structures connected in parallel with respect to acommon output line; and a signal line connected in parallel so that theplurality of digital control power source can communicate with oneanother, each of the operating digital control power sources outputtingcurrent value data of its own output current value converted into apulse width to the signal line and recognizing the current value datacorresponding to the maximum current value or the minimum current valuefrom the current value data of all of the operating digital controlpower sources as share current value data, and each of the operatingdigital control power sources controlling the output current on thebasis of the share current value data.

There is provided a device according to the seventh aspect, wherein eachof the digital control power sources has an open drain output terminal,and the open drain output terminals are connected in parallel to composewired OR.

There is provided a device according to the seventh aspect, whereinphases of output voltages of digital control power sources disagree withone another.

There is provided a device according to the seventh aspect, wherein anyone of the plurality of digital control power sources indicates phaseshift values of the respective output voltages to the other digitalcontrol power sources.

According to a eighth aspect of the invention, there is provided asemiconductor device comprising a digital PWM circuit outputting: afirst pulse signal switching from a low level to a high level at atiming of being synchronized with a clock signal and switching from thehigh level to the low level at a timing set by a shorter time resolutionthan that of a period of the clock signal; and a second pulse signalswitching from a low level to a high level at a timing set by a shortertime resolution than that of a period of the clock signal and switchingfrom the high level to the low level at a timing set by a shorter timeresolution than that of a period of the clock signal, the digital PWMcircuit including: a first delay circuit having a plurality of stages offirst delay elements connected serially; a first selection circuitselecting one from outputs of the respective delay elements andoutputting a signal setting a falling timing of the first pulse signal;a second delay circuit having a plurality of stages of second delayelements connected serially; a second selection circuit selecting onefrom outputs of the respective second delay elements and outputting asignal setting a rising timing of the second pulse signal; a third delaycircuit having a plurality of stages of third delay elements connectedserially; and a third selection circuit selecting one from outputs ofthe respective third delay elements and outputting a signal setting afalling timing of the second pulse signal, a delay amount in the seconddelay elements being longer than a delay amount in the first delayelements, and a delay amount in the third delay elements being longerthan a delay amount in the first delay elements.

There is provided a device according to the eighth aspect, wherein acontrol current supplied to the second delay elements is smaller than acontrol current supplied to the first delay elements, and a controlcurrent supplied to the second delay elements is smaller than a controlcurrent supplied to the third delay elements.

There is provided a device according to the eighth aspect, wherein thenumber of stages of second delay elements is smaller than the number ofstages of first delay elements, and the number of stages of third delayelements is smaller than the number of stages of first delay elements,

There is provided a device according to the eighth aspect, wherein anoutput of the first selection circuit is input to the first-stage seconddelay element in the second delay circuit.

According to a ninth aspect of the invention, there is provided asemiconductor device comprising a digital PWM circuit outputting: afirst pulse signal switching from a low level to a high level at atiming of being synchronized with a clock signal and switching from thehigh level to the low level at a timing set by a shorter time resolutionthan that of a period of the clock signal; and a second pulse signalswitching from a low level to a high level at a timing set by a shortertime resolution than that of a period of the clock signal and switchingfrom the high level to the low level at a timing set by a shorter timeresolution than that of a period of the clock signal, the digital PWMcircuit including: a first delay circuit having a plurality of stages offirst delay elements connected serially; a first selection circuitselecting one from outputs of the respective delay elements andoutputting a signal setting a falling timing of the first pulse signal;a second delay circuit having a plurality of stages of second delayelements connected serially; a second selection circuit selecting onefrom outputs of the respective second delay elements and outputting asignal setting a rising timing of the second pulse signal; a third delaycircuit having a plurality of stages of third delay elements connectedserially; and a third selection circuit selecting one from outputs ofthe respective third delay elements and outputting a signal setting afalling timing of the second pulse signal; an output of the firstselection circuit being input to the first-stage second delay element inthe second delay circuit.

There is provided a device according to the ninth aspect, wherein thefirst delay elements, the second delay elements, and the third delayelements have same structures.

1. A semiconductor device comprising: a plurality of digital controlpower sources comprising same structures connected in parallel withrespect to a common output line; and a signal line connected in parallelso that the plurality of digital control power sources can communicatewith one another, wherein each of the operating digital control powersources outputs current value data of its own output current valueconverted into a pulse width to the signal line and recognizes thecurrent value data corresponding to one current value from the currentvalue data of all of the operating digital control power sources asshare current value data through the signal line, and each of theoperating digital control power sources controls the output current onthe basis of the share current value data.
 2. The device according toclaim 1, wherein each of the digital control power sources comprises anopen drain output terminal, and the open drain output terminals areconnected in parallel to the signal line with wired OR.
 3. The deviceaccording to claim 1, wherein phases of output voltages of the digitalcontrol power sources disagree with one another.
 4. The device accordingto claim 1, wherein any one of the plurality of digital control powersources indicates phase shift values of the respective output voltagesto the other digital control power sources.
 5. The device according toclaim 1, wherein the one current value is a maximum current value fromthe current value data of all of the operating digital control powersources.
 6. The device according to claim 1, wherein the one currentvalue is a minimum current value from the current value data of all ofthe operating digital control power sources.